Advanced Differential Modulation Formats for Optical Access Networks

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JLT-14584-2012

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Advanced Differential Modulation Formats for Optical Access Networks Nikolaos Sotiropoulos, Graduate Student Member, IEEE, Ton Koonen, Fellow, IEEE, and Huug de Waardt, Member, IEEE  Abstract—The use of incoherent multilevel modulation formats with high spectral efficiency (more than two bits per symbol) has been proposed in order to enable the next generation of very high-speed Time-Division Multiplexing Passive Optical Networks (TDM-PONs). Incoherent multilevel modulation is attractive for access applications since multilevel formats allow the scaling of the bit rate with electronic and photonic components operating at a fraction of the bit rate. On the other hand, incoherent detection reduces the requirement for complicated Digital Signal Processing (DSP) and crucially an additional local oscillator, compared to coherent receivers. The modulation formats examined are Differential 8 Phase-Shift Keying (D8PSK) and three versions of incoherent 16 Quadrature Amplitude Modulation (QAM), specifically Star 16QAM, coded square 16QAM and 16QAM with pre-integration. Generation and detection of these formats is discussed, as well as the implementation challenges associated with such advanced modulation formats. The performance of these modulation formats was evaluated through extensive simulation and experimental work. Results indicate that incoherent modulation can fulfill important requirements of networks operators, namely increased bit rate and increased splitting ratio, and provide a cost-effective solution for Next-Generation Optical Access Networks. Index Terms—Passive optical network (PON), time-division multiple access (TDMA), multilevel modulation, differential detection.

I. INTRODUCTION

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HE explosive growth of Internet-based services, like file sharing, social networking, cloud computing and Internet video have fueled the demand for broadband access among home and business users. Forecasts indicate that these trends will only continue, with Internet video traffic rising at around 50% per year and file sharing at around 23% per year [1]. Gigabit passive optical networks have been widely deployed to satisfy this demand and 10 Gb/s versions have already been standardized ([2], [3]). However, even this increased capacity Manuscript received September 25, 2012. This work was supported by the Netherlands Ministry of Economic Affairs and the Netherlands Ministry of Education, Science and Culture through MEMPHIS Project. N. Sotiropoulos, A. M. J. Koonen and H. de Waardt are with COBRA Research Institute, Eindhoven University of Technology, Eindhoven, 5600 MB NL, The Netherlands (phone: +31402472091, e-mail: [email protected]). Copyright (c) 2013 IEEE.

might not be enough in the coming years [4]. In addition, new requirements from network operators, such as increased splitting ratio (more users served per PON) and longer reach, to enable a reduction in the number of Central Offices (CO) through node consolidation, and lower power consumption, have emerged [5]. At the same time, maintaining the cost-efficiency afforded by earlier PON installations is of paramount importance. To meet these requirements and enable cost-efficient highspeed broadband access networks, several network architectures and modulation formats have been proposed and investigated. Wavelength-Division Multiplexing (WDM) PONs have been extensively researched, since they offer logical point-to-point connections, utilize low-speed components and remove the need for bandwidth assignment protocols. On the downside, typically colored components are required in the Optical Network Unit (ONU) and port densities in the CO can be quite high. A number of solutions have been put forward to remove the need for colored components. Firstly, a tunable laser can be used in the ONU [6], but this is not a low-cost option for the time being. A second option is to provide unmodulated carriers from the CO [7] or remodulate the downstream signal [8]. In this option, Reflective Semiconductor Amplifiers (RSOAs) are typically used. However, the reach, splitting ratio and bit rate are limited due to Rayleigh back-scattering or residual modulation [9]. Furthermore, scaling the PON size can be problematic, due to the limited number of available wavelengths. Another approach utilizes coherent detection, which enables frequency selectivity by means of a tunable local oscillator and achieves superior sensitivity, making long-reach PONs without reach extenders possible [10]. Additionally, ultra-dense wavelength spacing is possible, due to the wavelength-selective nature of a coherent receiver with a tunable local oscillator. While such a scheme has very good performance, coherent receivers with tunable External Cavity Lasers (ECL), 900 hybrids, balanced detectors, Analog-to-Digital Converters (ADCs) and extensive DSP are currently not cost-efficient for access applications. Hybrid configurations, also known as WDM-TDM-PONs, stack a number of wavelengths, each of which serves a separate TDM-PON, typically at 10 Gb/s. This allows very high PON splitting ratios [9] without increasing the distribution loss linearly with the number of users. In addition, 10G transceivers specified in the PON standards can be

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JLT-14584-2012 utilized. On the other hand, the problem of wavelength selectivity in the ONU is as apparent as in WDM-PONs. An alternative to moving to WDM-based architectures is to increase the line rate of the TDM-PON to higher than 10 Gb/s, typically 40 Gb/s. The most straightforward way would be to employ 40 Gb/s On-Off Keying (OOK), but there are significant technical and cost challenges (need for 40 Gb/s burst-mode receiver, expensive 40 GHz electronics and photonics, receiver sensitivity issues) that make it unfeasible for access. One solution is to provide 40 Gb/s through four multiplexed 10 Gb/s OOK signals in different wavelengths [11], but the challenge of achieving a colorless ONU and the inefficient use of spectrum are significant drawbacks. A migration to multilevel modulation formats can remove the need for wavelength multiplexing, since high bit rates can be achieved while keeping the symbol rate at 10 GHz. A very popular approach to multilevel modulation is multi-carrier formats, such as Orthogonal Frequency Division Multiplexing (OFDM). 40 Gb/s OFDM-PONs have been presented, with both coherent [12] and direct-detection receivers [13]. For the coherent receiver case, very good receiver sensitivity is achieved, allowing high splitting ratios and long reach. As in the case of coherent WDM-PONs, the coherent receiver introduces considerable complexity in the ONU, which is undesirable. If direct-detection receivers are used, complexity is significantly reduced (in the optical domain), but receiver sensitivity suffers and splitting ratio and reach are severely limited. A different way to take advantage of the spectral efficiency of multilevel modulation is to utilize single-carrier, optical multilevel formats with incoherent (differential) detection. The differential receiver is similar to the coherent one, but the phase reference is provided by a delayed version of the signal itself and not by a laser source acting as a local oscillator. This also removes the need for carrier and phase recovery, greatly simplifying the required DSP. For certain formats and receiver architectures, DSP may be even fully discarded. The result is a less complex receiver, which however can provide better sensitivity than direct detection with a single photodiode at high bit rates. Where symmetric bit rates are required, a multilevel transmitter located in the ONU can provide the necessary bandwidth. Complexity can be reduced for the upstream channel by employing OOK modulation. A Directly Modulated Laser (DML) on a different wavelength than the downstream channel can then provide moderately asymmetric bit rate (e.g. 1:4 of the downstream bit rate) cost-efficiently. Optical multilevel modulation formats with differential detection have been extensively investigated, as a means to increase spectral efficiency in core and long haul networks. In that respect, the superior sensitivity of coherent detection, along with the possibility of mitigating impairments such as Chromatic Dispersion (CD) and performing polarization demultiplexing in the electronic domain- since the optical field is linearly mapped in the detected electrical field- has led to more deployments for core networks. Nevertheless, for access

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networks, where requirements are different, incoherent formats can be an attractive option. The first differential format that attracted attention was Differential Phase-Shift Keying (DPSK), where information is encoded in the phase difference between two consecutive symbols. Most popular implementations (with higher than a bit per symbol) involve Quadrature DPSK (DQPSK) [14] and D8PSK [15], encoding two and three bits per symbol, respectively. DSPK has attractive OSNR requirements and very good tolerance against nonlinearities [16], since it is a constant-amplitude format, for M  8 , but for higher phase levels the distance between symbols becomes so small that the trade-off between OSNR requirements and spectral efficiency becomes unfavorable. To further increase spectral efficiency without very high OSNR requirements, an amplitude component becomes necessary. There are several configurations of combined amplitude-differential phase formats. A straightforward way is to superpose Amplitude-Shift Keying (ASK) modulation over the DPSK signal. For example, if a binary ASK signal is superposed over a D8PSK signal, 16 Amplitude Differential Phase-Shift Keying (ADPSK), also known as Star 16QAM, signal is created, encoding four bits per symbol [17]. Similarly, a Quaternary ASK signal can be superposed to a DQPSK signal, also achieving four bits per symbol [18]. With this method, the amplitude and phase encoded bits can be encoded in the transmitter and decoded independently in the receiver. Depending on the transmitter and receiver structures used, DSP may not be necessary. If DSP-based transceivers are used, alternative 16-level formats are also possible, with arbitrary constellation diagrams. One approach uses a technique called phase pre-integration to encode the information in the amplitude and phase of the signal, which will result in a square 16QAM constellation diagram after detection and some (simple) DSP [19]. Alternatively, information can be encoded in such a way that transmitting square 16QAM constellation can enable correct decoding of the detected signals after a conventional differential receiver [20]. The paper is organized as follows. In Section II, the chosen modulation formats are described and suitable transmitter and receiver structures are discussed. In Section III, the proposed architecture of the PON is explained and the required subsystems are analyzed. Modeling of phase pre-integration and coded square QAM in VPI is shown in Section IV, results are presented and implementation challenges are discussed. In Section V, experiments with bidirectional transmission of D8PSK and Star 16QAM in a PON are described and their results are commented. Finally, on Section VI some conclusions are offered. II. INCOHERENT MULTILEVEL MODULATION FORMATS A. D8SPK and Star 16QAM D8PSK encodes three bits in the phase difference between two consecutive symbols, which belong to the



JLT-14584-2012 j

set e ,  { / 8,3 / 8,...(2M  1)   / 8} , where M=8 for this case. The resulting constellation diagram is shown in Fig.1a. The encoding procedure can be implemented either in DSP, if DSP-based transceivers are used, or by a discrete differential coder, if transceivers are driven by binary signals. The encoding amounts (functionally) to multiplying the symbol at moment k, ak , with the transmitted symbol at moment k-1,

sk 1 : sk  ak  sk 1 . For Star 16QAM, a fourth

independent bit adds amplitude modulation, creating two rings of D8PSK symbols, as it can be seen in the constellation diagram in Fig. 1b. This fourth bit does not need to be encoded and can be detected independently by an intensity detection branch of the incoherent receiver, in addition to the D8PSK detection part that remains unchanged. As previously mentioned, D8PSK/Star 16QAM signals can be generated either by binary signal-driven transmitters or by DSP-based transmitters (which utilize Digital-to-Analog Converters). In the first case, complexity is kept to a minimum in the electrical domain (only binary signals are required), but the optical part is complicated, requiring one modulator per bit. There are two configurations possible for binary-driven transmitters: the serial and the parallel configuration [16]. In the serial configuration, shown in Fig. 2a, after the data are deserialized and encoded, they drive three phase modulators cascaded in series, creating an 8-level phase modulated signal. The first phase modulator, responsible for creating the binary DPSΚ signal (0 and π shift), can be replaced by an amplitude modulator. For Star 16QAM, an amplitude modulator is added, also in series. In the parallel configuration, the two first phase modulators of the serial transmitter are substituted by a nested Mach-Zehnder modulator structure with a π/2 phase shift in the lower optical path (shown in Fig. 2b), known as an IQ modulator. Whilst the serial configuration is easier to control, since no DC biases are required (for pure phase modulation), the parallel configuration is preferable, because it is more immune to the amplitude noise of the driving signals. In a phase modulator, the amplitude ripples of the driving signal translate into phase ripples in the optical signal, whereas in an amplitude modulator (such as a Mach-Zehnder modulator) this is not the case [21]. The DSP-based transmitter is shown in Fig. 2c. The optical part consists of an IQ modulator, as the one described before. The driving signals for the IQ modulator are no longer binary, but multilevel. A D8PSK signal requires four-level electrical signals, while Star 16QAM requires eight-level signals. This transmitter configuration functions as a generic transmitter, since it can generate arbitrary optical signals, depending on the driving signals. In this scheme, complexity is moved on the electrical domain. The necessary driving signals are created in the DSP and are transferred to the analog domain through DACs. This allows the integration of functions in the DSP part, such as signal pre-distortion to account for the nonlinear characteristic of the MZ modulators and CD precompensation. In addition, flexible transmitters are made

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possible, changing the modulation format through software control, according to traffic requirements. For example, when bandwidth demand is low (e.g. at night), the modulation format can be switched from Star 16QAM to D8PSK, decreasing energy consumption on the ONU by shutting down the intensity detection part of the receiver. As with the transmitters, a number of different configurations exist for a multilevel differential receiver. Again, the trade-off is between complexity in the optical or the electrical domain. For binary detection schemes, M-DPSK formats require M/2 thresholds [22]. Translated to optical components, this requirement means that, for D8PSK, four Mach-Zehnder Delay Interferometers (MZDIs) with phase shifts of { / 8,3 / 8,5 / 8,7 / 8} and delay equal to one symbol period, each followed by Balanced Photo Detector (BPD) pairs are needed. Such a receiver is shown in Fig. 3a. Although this receiver allows for a very simple decoding scheme, it is very complex, power-inefficient and difficult to control, mainly because of the four phase shifters. The optical part of the receiver can be reduced to two MZDIs and BPDs, shown in Fig. 3b, if multilevel thresholding or analog electrical post-processing is employed [22], [23]. To accommodate a Star 16QAM signal, the receiver must include an intensity detection branch, which consists of a single photodiode, where part of the input signal is coupled. The beating of the two different intensity levels of the signal in the MZDIs, however, makes the decoding of the phase levels with multilevel thresholding difficult in the case of QAM signals. A generic, DSP-based differential receiver that can detect arbitrary differential multilevel formats is possible with some slight modifications of the receiver of Fig. 3b. The optical part of the receiver remains essentially the same, with the phase shifts being set at 0 and 900 on the upper and lower MZDI, respectively. The multilevel thresholding or the analog electrical processing network is removed. Instead, the detected electrical signals are sampled by ADCs and transferred to the DSP part of the receiver. Given that the incoming optical signal is E (t ) , the four outputs of the two MZDIs will be the following, not taking noise into account:

E1 (t )  E (t )  E (t  T ) E2 (t )  E (t )  E (t  T ) E3 (t )  E (t )  jE (t  T )

(1)

E4 (t )  E (t )  jE (t  T ) The balanced detection process will create two currents, I I , I Q (for the in-phase and quadrature component, respectively), which will be:

I I (t ) | E1 (t ) |2  | E2 (t ) |2 I Q (t ) | E3 (t ) |2  | E4 (t ) |2

(2)



JLT-14584-2012 After expanding the right-hand terms in (2), the resulting signal can be expressed as:

I I (t )  Re{E (t )  E  (t  T )} | E (t ) |  | E (t  T ) |  cos  IQ (t )  Im{E (t )  E  (t  T )} | E (t ) |  | E (t  T ) |  sin  ,

(3)

where  is the phase difference between the two consecutive symbols. Equation (3) shows that the resulting signal maps the real and imaginary part of the differential optical field, allowing the reconstruction of the original information symbol. That way, arbitrary constellation diagrams can be successfully detected and decoded. An equivalent version of (3), with the discrete notation used in the explanation of the encoding at the transmitter side, is:

I I ,k  Re{sk  sk1}  Re{(a k  sk 1 )  sk1}  Re{ak }  k 1

 k 1

I Q ,k  Im{sk  s }  Im{(a k  sk 1 )  s }  Im{ak },

(4)

from which it can be seen that the encoding is cancelled out and information is recovered. An alternative but mathematically identical implementation of the generic receiver involves a 2x4 900 hybrid, with a delay line in one of its inputs. This implementation is more suitable for receivers based on optical integration, since it leads to smaller devices and does not use controllable phase shifters. B. 16QAM with phase pre-integration D8PSK and Star 16QAM are the simplest formats possible for signals of high spectral efficiency, in terms of signal generation and detection. A drawback, however, of these modulation formats is that they are not optimal in the sense of minimum Euclidean distance between the symbols in the signal space. A more optimal packing of the symbols is achieved with square QAM constellations, which fall short 0.5 dB from the optimum arrangement [24]. The problem with square QAM constellations is that they don’t lend themselves easily to differential detection, because while the symbols in the inner and outer rings are evenly spaced (with π/4 spacing), as it can be seen in Fig. 4a, the symbols in the intermediate ring are asymmetric in phase. That means that the reconstructed differential signal, after the differential receiver, will not be a 16QAM signal. To enable differential detection of square 16QAM, the transmitted and received signals must be suitably processed. Such a scheme has been first proposed in [25], in the context of wireless communications, and it has been successfully implemented in optical communications in [19]. The scheme is based on a technique called phase preintegration, which amounts to adding the phase component of the previous transmitted symbol to the phase of the current one, while keeping its amplitude unchanged:

sk | ak |  exp( j  [arg(ak )  arg(sk 1 )])

(5)

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The resulting signal will have three magnitude levels, like square 16QAM, but it will have almost continuous phase distribution, due to the accumulation of phase after every symbol (Fig. 4b). In the receiver, the outputs of the BPDs are given by (3). After sampling and digitizing, the phase difference is estimated by computing the inverse tangent, with the two detected signals as arguments. The phase difference amounts to the phase of the original symbol ak , before the phase pre-integration operation in (5). A separate photodiode produces an estimate of the amplitude of the signal. The two components (phase and amplitude) are combined and the original square 16QAM constellation can be recovered. Due to the practically analog nature of the electrical driving and detected signals and the DSP required, only the generic, DSPbased transmitter and receiver described in the previous section can be used. C. Coded square 16QAM An alternative way of preserving the square 16QAM constellation is to encode the transmitted signal in such a way that the detector can unambiguously recover the original symbol from the outputs of the BPDs. This method requires a group of complex integers and a function F that ensures that the complex multiplication of any two symbols belonging to the group will be part of the group. This property of the function means that differential encoding and detection, two operations that involve complex multiplications, result in QAM symbols. F is a congruence operation that reduces the enlarged signal set that is the result of the multiplication of two QAM symbols into the original QAM set. This operation is essentially a Look-Up Table (LUT), where the product of the complex multiplication of two QAM symbols is unambiguously mapped into a third QAM symbol. It has been shown that a rotated by 450 16QAM signal set is such a group and a function as the one described exists for this set [20]. Note that the rotation of the constellation is necessary only during the signal processing. For transmission, the constellation can be rotated back to the conventional, since a phase shift common in consecutive symbols is cancelled out by the differential detection process. The encoding in the transmitter can be expressed as:

sk  F[ak  (sk1 )1 ]

(6)

In the receiver, the combination of the real and imaginary components, I I , I Q gives, as it has already been shown in (4), 

the complex multiplication sk  sk 1 . If this product is input to the mapping function, it follows by its properties [20] that:

F [ sk  sk1 ]  F [ F [ak  ( sk1 )1 ]  sk1 ]  F [ F [ak  [( sk1 )1  sk1 ]]  F [ak   ]  F [ak ]  ak

(7)



JLT-14584-2012 Thus, the original information symbol can be recovered from the differential constellation diagram (shown in Fig. 4c). Note that in coded square 16QAM, a separate intensity branch is not necessary for the detection of the signal; the differential metrics of (4) contain sufficient information to recover the QAM symbol. On the other hand, a DSP-based receiver is necessary, due to the need to implement a Look-Up Table (LUT) to map the received differential signal set to the square QAM signal set. For the generation of the square 16QAM signal, any of the many configurations proposed in the literature can be used. An extensive review can be found in [26]. The most straightforward solution, and the one typically used in long haul, high-order QAM with coherent detection experiments, is the IQ modulator. Four-level driving signals are required in this case. Multilevel signals can be created directly by DSP, where also the encoding takes place. Alternatively, four binary signals can be encoded by a discrete differential coder, as in D8PSK, and then can be electrically combined to create the two four-level signals that will drive the IQ modulator. This method of creating multilevel electrical signals, shown in Fig. 5, is widely used in high-speed QAM experiments [27]. III. PON ARCHITECTURE AND SUBSYSTEMS The motivation of employing differential multilevel modulation formats in access is to enable TDM-PONs with higher bit rate and splitting ratio, as stated in the Introduction. For that reason, the proposed network architecture is based on the typical TDM-PON configuration, seen in Fig. 6. In the CO, or Optical Line Terminal (OLT), as it is known in the PON nomenclature, the multilevel transmitter that generates the downstream signal and the receiver of the upstream channel are located. The two signals are separated by a circulator or a 1550/130 nm WDM diplexer, depending on the wavelength allocation scheme. Furthermore amplifiers may be employed, as a booster for the downstream signal and as pre-amplifiers for the upstream. The multilevel transmitter is more complex than transmitters used in current PONs, but since it is a shared resource among all the users in the PON, it is not so critical for the overall cost-efficiency of the network. The DSP-based configuration of Fig. 2c would be the optimal solution for the transmitter. This configuration is generic and can accommodate all discussed modulation formats and also compensate for chromatic dispersion, if it is required. Switching between formats of different spectral efficiency is also possible, as discussed before, adding more flexibility in the PON. In the Remote Node (RN), a passive 1xN coupler splits the downstream signal to all the PON users and combines the time-multiplexed upstream signals that will reach the OLT. Amplifiers, in particular Erbium-Doped Fiber Amplifiers (EDFAs), can also be located in the RN, if necessary. Traditionally, amplifiers are avoided in PONs, since they bring up costs. However, this begins to change, with EDFAs employed as boosters for the RF signal overlay in the XG-

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PON standard [28]. As the trend towards long-reach, highsplitting ratio PONs continues [9], [11], the utility of EDFAs as reach extenders is becoming apparent. With splitting ratios typically equal or higher than 256 users, the cost of EDFAs is spread among much more users than in legacy TDM-PON, which accommodate typically 32 or 64 users. The use of EDFAs can be additionally justified by the cost efficiency achieved by reducing the number of COs, made possible by such long-reach, high splitting ratio PONs. In the ONU, the downstream and upstream channels are again separated by a circulator or a diplexer, as in the CO. The ONU is the most cost-sensitive part of the network, since there is no cost sharing involved. For the upstream channel, there are two options: symmetric bit rate, which means mirroring the transmitter of CO in the ONU, and asymmetric bit rate, which utilizes simpler and more cost-efficient components. A multilevel transmitter in the ONU is not feasible in the near future, but it may be an attractive option when traffic demands reach the point where very high-speed upstream channels are necessary. In such a case, the required low-linewidth sources for the transmitter can be located in the ONU or can be centrally provided by the OLT. In the low-complexity, asymmetric option, the upstream channel is implemented by a DML operating at a different wavelength. The DML is a simple and cost-efficient solution that can provide bit rate up to 10 Gb/s and is compatible with existing PON standards. The channel bit rate is set to one fourth or one third of the downstream bit rate, which is moderately asymmetric, as is the case historically with most broadband technologies. In addition, note that emerging media services require extremely high downstream bit rates e.g. IPTV, High Definition TV (HDTV), Ultra-HDTV and 3DTV variants, hence symmetric bit rates are not required for the upstream. The differential receiver in itself is, however, quite complex, as described in the previous section. Photonic integration is instrumental in that respect. Integrated differential receivers on silicon have already been demonstrated with very good results [29]. Since access networks are a mass-volume market, it is conceivable that such devices can become cost-efficient, if significant market adoption is achieved. As with the transmitter, the versatility of the DSP-based configuration is preferable also for the receiver. This configuration allows the evaluation of all proposed modulation formats, while minimizing the optical front-end. An important issue for the DSP-based receivers is the sensitivity, power consumption of the DSP chipsets and ultimately cost. Due to extensive efforts for the development of coherent, QPSKPOLMUX based 100 Gb/s transponders for long-haul applications, the state-of-the-art commercial ADCs have sample rates in excess of 50 GSa/s [30]. For access application, a sampling rate of 20 GSa/s (up sampling by a factor of two for 10 GHz signals) is well within current capabilities. High-speed ADCs are increasingly being demonstrated with CMOS technology allowing single-chip ADC and DSP solutions [31]. This configuration can enable



JLT-14584-2012 lower power consumption, better yield compared to other approaches, e.g. SiGe ADCs and Si DSP, and of course the proven cost-efficiency benefits of CMOS technology [32]. CMOS Application-Specific Integrated Circuits’ (ASICs) costs are volume-driven, when based on mature technologies. Again, the relatively low speed of access applications (compared to long haul) means no cutting-edge designs and exotic materials are required and the potentially large volumes suggest that ASICs for PONs fit well into this pricing model. An important factor that determines the development effort required for the ASIC, as well as its power consumption, is the complexity of the signal-processing algorithms [33]. In that respect, incoherent detection has a clear advantage over coherent detection and OFDM. All DSP-based receivers share some common functions, such as de-skew, timing recovery, normalization and symbol estimation and decoding [34]. Besides these common elements, coherent receivers require frequency and phase recovery. CD compensation is also typically implemented. For OFDM receivers, the more computationally intensive task is the Fast Fourier Transform (FFT). Other tasks involve removal of the Cyclic Prefix and frequency domain equalization. Apart from the common algorithms, incoherent receivers for D8PSK and Star 16QAM do not require any further DSP. The QAM with phase preintegration requires-besides the common algorithms described above- the computation of the inverse tangent and the reconstruction of the amplitude and phase component of the signal. For the coded square QAM, an implementation of a LUT that maps the differential signal set to the original square QAM set is the only extra functionality needed. It is clear that the development effort and the power consumption, which is a significant part of operational costs, of the incoherent receivers can be kept at significantly lower levels compared to the other multilevel approaches. The proposed TDM-PON architecture can be combined with WDM techniques (as the hybrid WDM-TDM PON proposed in [9]) to further scale the PON size and facilitate the convergence of metro and access networks. The wavelength stacking of independent TDM-PONs has the further advantage of enabling cost sharing of components (such as optical amplifiers) among an even larger number of users. Additionally, the increased bit rate of the multilevel TDMPON enables significant increase on the sustainable bit rate per user, that for 10 Gb/s PONs with splitting ratios higher than 128 users suffers. IV. MODELING AND SIMULATION RESULTS To evaluate the more sophisticated multilevel formats presented, 16QAM with pre-integration and coded square 16QAM, a TDM-PON with the respective transceivers was modeled in VPI Optical Systems simulation software. With this approach, the performance of the modulation formats could be examined, since analytical formulation of the error probabilities is difficult, due to the unconventional constellation diagrams and nonlinear operations in the

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receiver.1 A. Symmetrical 10 Gb/s 16QAM with phase pre-integration simulations The PON model used in the simulations is shown in Fig. 7 [35]. A symmetric configuration, with 10 Gb/s 16QAM signals for both downstream and upstream channels and identical transceivers in the CO and ONUs, is chosen. The lowlinewidth (200 kHz, as for the downstream signal) Continuous Wave (CW) light for the upstream channel is generated centrally in the CO and is distributed to the ONUs, spaced 3.2 nm away from the downstream wavelength at 1550 nm. This removes the need for an expensive ECL in every ONU, utilizing a single source for the entire PON. WDM (de)multiplexers are included in the model, to accommodate the possibility of stacking a number of TDM-PONs on different wavelength pairs. It was not possible to include, however, these WDM channels in the simulation, due to computational constraints of bidirectional propagation over the fiber model in VPI. A splitting ratio of 32 users is assumed in the model. The amplification scheme of the PON is based on Semiconductor Optical Amplifiers (SOAs). The SOAs are located in the ONU, where they amplify the downstream signals, as well as the CW carrier, and in the CO, where they are used as pre-amplifiers for the upstream signal. Optical amplification is generally not preferable in the ONU, but SOAs can be integrated in the receiver design and thus do not introduce unreasonable complexity. Regarding the effects and impairments modeled, all optical impairments are included in the simulations. The generation and sampling of the multilevel electrical signals is noise and jitter free, to keep focus in the Amplifier Spontaneous Emission (ASE) noise-limited regime. The BER versus OSNR curves have been obtained experimentally in [19], but in access networks receiver sensitivity is a more important metric, since it determines the power budget of the network and therefore the maximum splitting ratio and reach. The first aspect of the system design that was investigated was the optimal power of the unmodulated carrier. As the upstream signal is more sensitive than the downstream, the carrier power was optimized for the upstream transmission. With a constant downstream input power of -8.6 dBm at the fiber, resulting to a BER for the downstream channel of around 10-11, the upstream BER as a function of the carrier input power is shown in Fig. 8. The increase of the carrier power leads to improved BER for the upstream signal, until the nonlinear effects start to degrade the signal at around 1.5 dBm input power. The lowest BER occurs for 0 dBm input power and is 5‧ 10-7, without Forward Error Correcting (FEC) codes. Using the above specified values, the BER as a function of the received power was evaluated. Since the transmitter and the receiver were custom-made (not part of the VPI library), it



JLT-14584-2012 was not possible to use the BER estimation tools of VPI and error counting leads to prohibitive simulation times. The approach followed was to use the Error Vector Magnitude (EVM) of the received signal as an estimation of its SNR, and use the SNR to obtain a BER value assuming a Gaussian distribution of the noise. In reality, the noise is not Gaussian, as it can be observed in the received constellation diagrams (Fig. 9), but it affects more intensely the phase of the received signal. This is a common characteristic of all systems employing both interferometric and intensity detection. The differential phase component is corrupted by noise from two observation instances, doubling its noise variance, while the amplitude component is affected by noise from a single instance. This non-Gaussian noise distribution leads to an underestimation of the error probability, if conventional, Euclidean boundary metrics are implemented. However, if non-Euclidean metrics for symbol detection are used, taking into account the increased noise in the radial (phase) direction, improved BER performance is achieved [19], justifying the approximation. The BER curves for back-to-back and 20 km of fiber are shown in Fig. 10. The power penalty for the downstream signal is negligible. For the upstream signal, however, there is larger power penalty. It is mainly caused by the nonlinear interactions in the fiber, with SPM and XPM being the most important, as the signal is phase-modulated. Further simulations, with the nonlinear effects turned off, showed that the power penalty becomes negligible in that case. Besides the penalty because of the fiber transmission, a significant degradation of the performance for the upstream signal is observed, compared to the downstream. The reason for this degradation is the second amplifier that lies in the upstream path. When the noise from the preamplifier at the CO is turned off, the BER of the upstream signal is comparable to that of the downstream. However, even with this sensitivity degradation, error-free operation for the upstream signal can be achieved, as the BER can be well below the FEC limit, which is around 10-3, for an overhead of 7%. B. Symmetrical 10 and 40 Gb/s coded square 16QAM simulations The PON model used in the simulations with the coded square 16QAM can be seen in Fig. 11 [36]. The PON offers symmetric bandwidth, with simulations done for bit rates of 10 and 40 Gb/s, over 20 km of SSMF. The CW carrier is again provided by the CO, to keep the ONU source-free. The main difference with the network architecture of the pre-integration QAM simulations is that optical amplification is moved to the RN and the (de)multiplexers (for WDM stacking) are not included. Two EDFAs, located before the splitter, amplify the downstream channel and the unmodulated carrier, and the upstream channel, respectively. Another EDFA is used as a pre-amplifier in the CO. The transfer of optical amplification 1 A theoretical calculation of the BER performance of phase preintegration QAM was carried out in [25], but a coherent receiver was assumed and the phase of each symbol was estimated individually.

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in the RN further simplifies the ONU hardware and shares the cost among all the users, since the amplifiers are placed before the splitter. Since there is no need for integration in this amplification scheme, EDFAs, which have a lower noise figure than SOAs, are used. A more realistic modeling of the driving signals is implemented, including noise and filtering. The Bit Error Rate (BER) curves as a function of the OSNR for 10 and 40 Gb/s downstream and upstream transmission are shown in Fig. 12 and 13. For the upstream case, coherent detection was also included in the simulations, for comparison reasons. Carrier and phase recovery was implemented in the coherent receiver, using an algorithm provided in VPI, but no CD compensation was performed. In the CO, the receiver is shared by a number of users (32 in this case), so a more expensive solution can be justified. The linewidth of the lasers involved was 100 kHz and fiber input powers were 0 dBm for the carrier and -4 dBm for the downstream signal. The received power was -6 dBm (right before the receiver) for both channels. The results indicate that for the downstream channel, error-free (with FEC) transmission can be achieved for both 10 and 40 Gb/s. In the upstream channel, error-free incoherent detection is possible for 10 Gb/s, but for 40 Gb/s there could be no errorfree operation within the OSNR limits of the amplification scheme. If coherent detection is used, error-free operation is possible for 10 and 40 Gb/s. For incoherent receivers, BER was obtained by direct error-counting. For the coherent receivers, BER was estimated using VPI’s built-in tools. C. Simulation results discussion and implementation challenges Simulations showed that the DSP-based, square QAM modulation formats can be attractive options for future TDMPONs. 10 Gb/s symmetric operation over 20 km fiber and at least 32 users is shown for both formats. For the 40 Gb/s coded 16QAM PON, error-free downstream transmission is possible. For the upstream channel, coherent detection can provide the required sensitivity or a lower bit rate signal can be used. For the QAM with pre-integration, significant margin is available, which indicates that error-free operation at 40 Gb/s with FEC and a high splitting ratio are possible. Regarding OSNR requirements, the pre-integration technique offers superior performance [19], but on the other hand, the simpler receiver and processing of the coded QAM can be advantageous in access applications. To enable comparison of the proposed advanced modulation formats with conventional OOK, the receiver sensitivities for 10 Gb/s OOK that are

specified in the relevant standards, along with the sensitivities obtained through simulations and experimental work are shown in Table 1. Comparing the obtained results with OOK-based systems, for the 10 Gb/s phase pre-integration 16QAM it can be seen that considerably higher splitting ratio and longer reach can be achieved, due to the format’s high sensitivity, while operating at one fourth of the bit rate. An implementation power penalty, for reasons



JLT-14584-2012 extensively discussed in the following paragraphs, will reduce the receiver sensitivity, but it can be compensated for by an increased input power to maintain the high power budget. For the coded square QAM, receiver sensitivity and as a result power budget has not been estimated, but due to smaller symbol spacing it is expected to be lower than the phase preintegration results. The main motivation for the simulation work on these advanced modulation formats was to identify the theoretical limits of the sensitivity of the incoherent receivers, assuming perfect driving signals, modulation and detection processes. For an experimental implementation of the systems explored in simulations, a number of non-idealities occur. The first issue is the generation of the multilevel signals. For the signals required for these two formats, the only option for laboratory demonstrations is a high-speed Arbitrary Waveform Generator (AWG). However, the signal quality of the AWG is not (yet) comparable with the quality achieved in Binary Pattern Generators (BPGs). A four-level signal (that generates the I or Q component of a square 16QAM signal) from the available AWG, at 2.5 GHz, is shown in Fig. 14. There are considerable ringing oscillations in the signal. Taking into account that for the phase pre-integration technique, the driving signals have tens of levels, it is obvious that there is significant degradation compared to the ideal case in simulation work. A further related problem is the linearity of the electrical amplifiers, both of the drivers in the transmitter side and of the TransImpedance Amplifiers (TIAs) and pre-amplifiers in the receiver side. As the dominant modulation format in highspeed optical communication systems so far has been OOK, amplifiers are optimized for binary signals. The introduction of commercial (D)QPSK transceivers has not changed the situation, since they also operate with binary signals. This means that the amplifiers are another source of distortion for multilevel signals. The introduction of coherent 16QAM transceivers requires improvements in this respect and will lead to the availability of suitable linear amplifiers. It should be noted that these kinds of distortions are not easy to be modeled in VPI, so their impact cannot be estimated before the experimental validation. The sensitivity of the advanced modulation formats under discussion to deviations from the optimal operating points of the transmitter and the receiver is also a source of potential degradation in system performance. Deviation from the optimal amplitude of the driving signals, the proper biasing of the IQ modulator, the optimal phase shifts in the MZDIs and imbalances in the receiver can affect signal quality and receiver performance. The best signal quality is achieved when the driving signals have amplitude of 2  V , where V is the voltage required for an 1800 phase shift in the optical signal. For lower driving voltages, the signal is distorted. For example, the simulated received constellation diagrams for the coded 16QAM format for signals with amplitude at the optimal value and at 2/3 of it are shown in Fig. 15 a and b respectively, where the reduction of the inter-symbol distances

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for the reduced amplitude case can be clearly observed. The biasing of the two MZ modulators of the IQ superstructure in the ‘null’ transmission point and the biasing of the phase shifter in the quadrature point are also critical. After initial setting of the DC biases, environmental changes can lead to drifting and consequently to signal degradation. Fig. 15c shows the constellation diagram when both lower driving voltage (2/3 of the optimal, as before) and a 100 deviation for the phase shifter are simulated. For commercial implementations, the operating points for the DC biases can be maintained with control circuits using feedback from photodiodes located in the modulator structures. In the receiver side, the possible impairments include deviations from the optimal values of the phase shifters in the MZDIs (or the 900 hybrid, depending on the implementation), deviations from the nominal delay, non-ideal extinction ratio of the MZDI and amplitude and timing mismatch in the BPD. It has been shown for DQPSK in [37] that the last four have small impact on the achievable BER for reasonable deviations. Similar conclusions have been reached for 16QAM with phase pre-integration in [38]. On the other hand, the phase detuning can be detrimental for the performance of the system, if left uncompensated, because it results in a rotated constellation diagram. It is possible, however, to recover the correct alignment of the in-phase and quadrature components through orthonormalization algorithms in the DSP, such as GramSchmidt or Löwdin [34]. V. EXPERIMENTAL WORK AND RESULTS For the reasons described extensively in the previous section, experimental work on PONs with differential multilevel formats was focused on D8PSK and Star 16QAM. These two modulation formats put considerably less strain in electrical components (AWG and amplifiers) when using multilevel signals and can also be generated by binary driving signals, trading complexity on the optical domain with simpler electronics. A. 10 Gb/s Star 16QAM/OOK PON The experimental set-up of a TDM-PON based on a 10 Gb/s, DSP-based Star 16QAM downstream channel and 2.5 Gb/s OOK upstream channel is shown in Fig. 16 [39]. The differential encoding of the downstream data, as well as the pre-distortion to compensate for the IQ modulator’s non-linear transfer function, is performed offline in Matlab and the resulting waveforms are loaded to the AWG. The transmitted sequence is 100k symbols long and the ECL source operates at 1550 nm, with a linewidth of 100 kHz. The created Star 16QAM signal is then launched from the Central Office (CO) unamplified into 25 km of SSMF through a circulator, with an input power of -4.5 dBm. The RN consists of two circulators that separate the downstream and the upstream channel, two EDFAs and an attenuator that introduces 15 dB attenuation, modeling the loss of 1:32 splitter. The EDFA amplifies the downstream signal from -12 dBm to +10 dBm. In the ONU, an EDFA is used as a pre-amplifier to boost the received power to



JLT-14584-2012 7 dBm, followed by a filter to remove excess ASE noise. Preamplification is necessary in this experiment, since the BPDs used do not have integrated TIAs and their output voltage is very low. BPDs with integrated TIAs are commercially available, but typically with a limiting amplifier configuration. This configuration is optimal for DPSK and DQSPK, which produce binary signals, but is detrimental when multilevel signals need to be amplified. A portion of the input power, through a 70:30 splitter, is sent to a single photodiode for amplitude detection. The electrical signals are amplified (except for the amplitude component), sampled by a Digital Phosphor Oscilloscope (DPO) at 50 GSa/s and transferred to Matlab for offline processing. The 2.5 Gb/s NRZ driving electrical signal for the upstream channel is created by a 12.5 Gb/s Bit Pattern Generator (BPG). It then modulates a DML in the ONU, which has an output power of 0 dBm and operates at 1554 nm. The signal is introduced through a circulator into the drop fiber and then into the RN, where it is attenuated by the splitter and forwarded into the upstream-path EDFA. The signal is then amplified to +3 dBm (minimum output power of the EDFA) and then attenuated, so it does not affect the downstream signal. The EDFA in the upstream path was necessary in this experiment, as the sensitivity of the PIN-TIA receiver used is 23 dBm, while the total losses in the system are 27 dB. However, if a better receiver (e.g. with an APD) or a transmitter with higher output power is used, the EDFA in the upstream channel may be removed. Detected constellation diagrams of the Star 16QAM signal for back to back and transmission over 25 km of fiber are shown in Fig. 17. The 16 signal points are clearly separated and the predominance of phase noise can be observed. The differential phase component is corrupted by noise from two observation instances (as it depends on the current as well as the previous symbol), doubling its noise variance, while the amplitude component is affected by noise from a single instance. From the constellation, it is obvious that an important parameter in the system design is the radius ratio between the two D8PSK circles. Since phase noise is twice as large as amplitude noise, the inner circle should have as large radius as possible, to exploit the larger distance between its symbols, which is proportional to its radius. However, if the two circles are spaced too close to each other, amplitude errors begin to dominate. To find the optimal operating conditions, the BER of the Star 16QAM signal as function of the radius ratio has been evaluated and the results are shown in Fig. 18. The minimum BER is found for values between 0.65 and 0.75 (normalized to the outer ring radius of 1), so the inner ring radius was chosen to be 0.7. The BER of the Star 16QAM signal as a function of the input power at the receiver was measured by tuning the optical attenuator before the EDFA. The results can be seen in Fig. 19. Error-free transmission with FEC, introducing 7% overhead, is possible. The sensitivity for the FEC threshold of 10-3 is around -31 dBm, which indicates very good transmitted

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signal quality and detection process. There is negligible power penalty observed due to fiber transmission. The corresponding BER curve for the upstream OOK channel is shown in Fig. 20. The sensitivity of the signal at BER of 10-9 is around -22 dBm. Very small power penalty due to fiber transmission is observed and the sensitivity measured is quite close to the reference sensitivity of the particular receiver, indicating minimal interference of the downstream channel. B. 30 Gb/s D8PSK/OOK PON For the very-high speed PON experiment [40], with a symbol rate of 10 GSa/s, binary-signal generated D8PSK was chosen. The available AWG has a bandwidth of around 6 GHz, so the DSP-based approach was not possible. Again, a DML provides an OOK upstream signal, this time at 10 Gb/s and at 1310 nm. The experimental set-up can be seen in Fig. 21. In the CO, a BPG creates the binary signals with bit pattern of 213-1. The signals are decorrelated through different cable lengths, aligned and amplified, and then drive the D8PSK serial modulator. An ECL set at 1550 nm functions as the lowlinewidth CW light source. The D8PSK signal passes through 4km of Dispersion Compensating Fiber (DCF) that compensates for the dispersion over the transmission fiber and is then amplified. After a 1550/1310 nm diplexer, the signal is launched into 25 km of SSMF with an input power of around 0 dBm. The RN is entirely passive in this case. An attenuator introduces 15 dB of splitting losses (that corresponds to a splitting ratio of 32 users). In the ONU, the downstream signal is separated through a diplexer and is pre-amplified, before being detected and sampled by a Digital Phosphor Oscilloscope (DPO), with a captured length of 100k symbols. Optical pre-amplification was again necessary, as the same BPDs without TIAs as in the Star 16AM experiment were used. Detected constellation diagrams for back to back and transmission over 25 km of fiber are shown in Fig. 22a and b. No degradations from imperfect generation, detection processes or fiber transmission are observed. For the upstream channel, a 12.5 Gb/s BPG creates a 10 Gb/s NRZ signal with a pattern of 223-1 that drives the DML. The OOK upstream signal, after passing through the splitter and 25 km of fiber, is pre-amplified in the CO by an SOA to boost the signal before being detected by a PIN-TIA receiver, which has a nominal sensitivity of -19 dBm. If a more sensitive APD-TIA were available at the CO, the SOA would be unnecessary. Detected OOK eyes for back to back and transmission over 25 km of fiber are shown in Fig. 23a and b, also exhibiting no sign of degradation. The BER curves as a function of the received power for bidirectional transmission of the downstream and upstream channels can be seen in Fig. 24 and Fig. 25. For the D8PSK signal, error-free transmission with FEC, introducing 7% overhead, is possible. Observed sensitivity is -33.5 dBm and a small power penalty less than 1 dB is measured. The results validate the very good generation and detection of the multilevel signal. For the OOK upstream channel, receiver sensitivity is -24 dBm, with very low power penalty due to fiber transmission and no error floor. If a commercial APD



JLT-14584-2012 receiver is used, improved receiver sensitivity can be achieved. C. Discussion Experimental results show that very good receiver sensitivity is achieved for both D8PSK and Star 16QAM, for the preamplified receiver. For the Star 16QAM experiment, it is shown that, for conventional fiber lengths, there is no need for amplification in the RN. Comparing to the 10 Gb/s OOK standards (Table 1), superior sensitivity is achieved, in addition to increased spectral efficiency (by a factor of four). The available power budget is 31 dB, assuming 0 dBm input power, which is in line with 10 Gb/s standards [2], [3]. For a fiber length of 25 km, up to 256 users can be accommodated. If both long reach and large splitting ratio are required, the EDFA in the RN can function as a reach extender, increasing the available power budget. By moving to lower-order modulation (D8PSK) and binary-signal generation, even higher sensitivity is obtained, despite the fourfold increase in the symbol rate. As a result, for the D8PSK experiment the power budget is even higher, around 34 dB (assuming 0.5 dBm input power). This result represents a 3 dB increase over existing PON standards for comparable input powers, while providing three times higher bit rate in the downstream channel. This extra power budget can be translated into 15 km longer reach or double splitting ratio and clearly showcases the potential of advanced modulation formats as an enabling technology for future PONs. This value allows a splitting ratio of 512 users, again for 25 km-long PONs. Alternative allocations of the budget are naturally possible, e.g. 60 km fiber length and 128 users. Even longer distances and higher PON sizes are possible, if an in-line amplification scheme, like the one used in the Star 16QAM experiment, is implemented. Regarding power penalties, no degradation due to the bidirectional transmission over a single fiber with the upstream channel is observed, as well as no impact of Rayleigh backscattering. In general, the suitability of the differential formats for access applications has been demonstrated and the performance of the generic, DSP-based receiver has been proved. The splitting ratios and fiber lengths afforded by the achieved power budgets enable the reduction of the number of COs required and the upgraded system rate allows increased available bandwidth per user. The subsequent reduction of capital and operating expenses for the operators can justify the introduction of more complex, DSP-based transceivers in access networks. The complexity of the set-ups used for the experimental proof-of-concept can be considerably reduced, making the proposed systems more suitable for future commercial deployments. DCF can be removed, if DSP-based transmitters, which enable CD compensation by pre-distorting the signal before transmission, are utilized. CD compensation can also be performed in the receiver side [41]; however it is not preferable since it introduces additional complexity in the ONU. In actual PON deployments, it is not desirable to employ optical pre-amplification in the ONU. However, if balanced photo detectors with linear TIAs are used, un-

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amplified receivers with satisfactory sensitivity will be feasible. Any power penalty due to the lack of optical preamplification can be offset by increasing the transmitted power. Another option is to perform single-ended detection on each output of the demodulator with PIN/TIA, as photodiodes with linear TIAs are available, and perform the balancing in the digital domain, after sampling. Theoretically, the two aforementioned detection schemes have the same performance, but the limited resolution of ADCs means that balanced detection is preferable in practice [34]. The proposed system can be further simplified by employing more cost-efficient laser sources, such as DFB lasers. It has been shown [42] that a linewidth of around 1.5 MHz (at 10 Gsym/s) incurs modest penalty of 1 dB for D8PSK, for BER close to the FEC limit. Regarding the required ADC rates, for 10 Gsym/s signals a sampling rate of 20 Gsa/s is sufficient for the reconstruction of the signal, which can ease the requirements for high-speed electronics in the ONU. VI. CONCLUSION In this paper, a review of our proposals to use differential multilevel modulation formats in access networks is given. It has been shown through simulations and experimental results that differential formats with high spectral efficiency can enable Next-Generation PONs, with increased bit rate (30 Gb/s shown experimentally), splitting ratio (up to 512 users) and potentially reach. These results represent the highest combination of system bit rate and splitting ratio achieved so far for PONs without coherent receivers on a single wavelength and polarization. The cost aspects of such systems has been explored, with the analysis showing that the progress in optical integration and silicon technology can allow costefficient receivers, if mass-market criteria are met. Implementation challenges have been discussed and the bottlenecks in terms of components for the introduction of advanced, DSP-based incoherent 16QAM formats have been identified. The presented results indicate a promising future for competitively priced incoherent receiver based ONUs to support high spectral efficiency in access networks. REFERENCES [1] [2] [3]

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[28] S. Jain, F. Effenberger, A. Szabo, Feng Zhishan, A. Forcucci, Guo Wei, Luo Yuanqiu, R. Mapes, Zhang Yixin, V. O'Byrne, "World's First XGPON Field Trial," J. Lightw. Technol., vol.29, no.4, pp.524-528, Feb.15, 2011. [29] C.R. Doerr, L. Chen, “Monolithic PDM-DQPSK receiver in silicon,” presented at Europ. Conf. on Opt. Commun. (ECOC), 2010, paper PDP3.6. [30] N.Cvijetic, "OFDM for Next-Generation Optical Access Networks," J. Lightw. Technol., vol.30, no.4, pp.384-398, Feb.15, 2012. [31] I.Dedic, “High-speed CMOS DSP and data converters,” presented at Optical Fiber Conf. (OFC), 2011, paper OTuN1. [32] I.Dedic, "56Gs/s ADC: Enabling 100GbE," presented at Optical Fiber Conf. (OFC), 2010, paper OThT6. [33] S.J. Savory, “Digital Signal Processing Options in Long Haul Transmission”, presented at Optical Fiber Conf. (OFC), 2008, paper OTuO3. [34] S.J. Savory, "Digital Coherent Optical Receivers: Algorithms and Subsystems," IEEE J. Sel. Topics in Quantum Electron., vol.16, no.5, pp.1164-1179, Sept.-Oct. 2010. [35] N. Sotiropoulos, T. Koonen, H. de Waardt, "Bidirectional incoherent 16QAM transmission over hybrid WDM/TDM passive optical network," presented at 12th Intern. Conf. on Transparent Opt. Netw. (ICTON), 2010, paper Th.A2.4. [36] N. Sotiropoulos, H. de Waardt, T. Koonen, "Novel 16QAM Detection Scheme for Optical Access networks," presented at Access Netw. and In-house Comm., Toronto, Canada, paper AMC5, 2011. [37] G. Bosco, P. Poggiolini, "On the joint effect of receiver impairments on direct-detection DQPSK systems," J. of Lightw. Technol., vol.24, no.3, pp.1323-1333, March 2006. [38] N. Sotiropoulos, A.M.J. Koonen, H. de Waardt, ”On the effect of receiver impairments on incoherent QAM systems,” Presented at the 15th Annual Symposium of the IEEE Photonics Benelux Chapter, 2010, Delft, The Netherlands, pp. 29-32. [39] N. Sotiropoulos, A.M.J. Koonen, H. de Waardt, “Star 16QAM/OOK Bidirectional Transmission over a TDM-PON”, in Microw. and Opt. Technol. Lett., accepted for publication [40] N. Sotiropoulos, A.M.J. Koonen, H. de Waardt, ”Experimental Demonstration of an Incoherent TDM-PON with 30 Gb/s D8PSK Downstream and 10 Gb/s OOK Upstream Data,” presented at Europ. Conf. on Opt. Commun. (ECOC), 2012, paper Tu.3.B.2. [41] X. Liu, X. Wei, "Electronic Dispersion Compensation Based on Optical Field Reconstruction with Orthogonal Differential Direct-Detection and Digital Signal Processing," presented at Optical Fiber Conf. (OFC), 2007, paper OTuA6. [42] Y. Han, and G. Li, "Sensitivity limits and degradations in OD8PSK," IEEE Photon. Technol. Lett., vol.17, no.3, pp.720-722, March 2005.



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Table 1 Modulation formats and receiver sensitivity

Modulation format/ bit rate OOK/ 10 Gb/s (XG-PON) [2] OOK/ 10 Gb/s (10G-EPON) [3] Phase pre-integration 16QAM/ 10 Gb/s Star 16QAM/ 10Gb/s D8PSK/ 30 Gb/s

Receiver sensitivity (dBm) -22.5 (PIN)/ -28 (APD) -20.5 (PIN)/ -28.5 (APD) -39 (pre-amplifier) -31 (pre-amplifier, no TIA) -33.5 (pre-amplifier, no TIA)

Fig. 1a D8PSK constellation diagram

Fig. 1b Star 16QAM constellation diagram

Fig. 2a Serial transmitter



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Fig. 2b Parallel transmitter

Fig. 2c DSP-based transmitter

Fig. 3a Binary differential receiver

Fig. 3b Multilevel differential receiver



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Fig. 4a 16QAM constellation diagram

Fig. 4b Transmitted constellation diagram with phase pre-integration

Fig. 4c Differential constellation diagram, coded 16QAM

Fig. 5 Binary square 16QAM transmitter



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Fig. 6 TDM-PON architecture

Fig. 7 Phase pre-integration 16QAM simulation model

Fig. 8 Upstream channel BER vs. carrier input power

Fig. 9 Phase pre-integration 16QAM received constellation diagram



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Fig. 10 BER vs. received power, phase pre-integration 16QAM PON

Fig. 11 Coded 16QAM simulation model

Fig.12 BER vs. OSNR downstream channel, coded 16QAM PON

Fig. 13 BER vs. OSNR upstream channel, coded 16QAM PON

Fig.14 4-level driving signal at the output of the AWG at 2.5 Gsym/s



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Fig. 15a: Simulated constellation diagram coded 16QAM, ideal case

Fig. 15b Simulated constellation diagram coded 16QAM, 2/3 driving voltage

Fig. 15c Simulated constellation diagram coded 16QAM, 2/3 driving voltage and 80 0 phase shift

Fig. 16 Experimental set-up, Star 16QAM/OOK PON



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Fig. 17 Star 16QAM received constellation diagrams (a) back to back (b) 25 km fiber

Fig. 18 BER vs. inner ring radius (normalized to 1 for outer ring)

Fig. 19 BER vs. received power, 10 Gb/s Star 16QAM

Figure 20 BER vs. received power, 2.5 Gb/s OOK



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Fig. 21 Experimental set-up, D8PSK/OOK PON

Fig. 22 D8PSK received constellation diagrams (a) back to back (b) 25 km fiber

Fig. 23 OOK received eyes (a) back to back (b) 25 km fiber

Fig. 24 BER vs. received power, 30 Gb/s D8PSK



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Fig. 25 BER vs. received power, 10 Gb/s OOK


Advanced Differential Modulation Formats for Optical Access Networks

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