Antennas for Wireless Communications – Ba Basi sic c Princ Princip iple les s and and System Applications Warren Stutzman and Bill Davis Virginia Tech Antenna Group June 9, 2006 antenna.ece.vt.edu
OUTLINE 1. Introduction 2. Antenna Fundamentals 3. Antenna Elements 4. Array Antennas
Stutzman Davis Davis Stutzman
break
5. System Considerations 6. Wireless Applications 7. Photo presentat presentation ion
Davis Both Stutzman
1. INTRODUCTION • • • • • •
The Speakers Self Introductions of Class Wireless versus Wireline History of Communications The Spectrum Antenna Performance Parameters
Warren Stutzman • Professor at VT for 37 years • Fellow of the IEEE • Past President of IEEE Ant & Prop. Society • Distinguished alumnus of U of Illinois • Founder of Virginia Tech Antenna Group • Served as ECE Dept Head twice
1. INTRODUCTION • • • • • •
The Speakers Self Introductions of Class Wireless versus Wireline History of Communications The Spectrum Antenna Performance Parameters
Warren Stutzman • Professor at VT for 37 years • Fellow of the IEEE • Past President of IEEE Ant & Prop. Society • Distinguished alumnus of U of Illinois • Founder of Virginia Tech Antenna Group • Served as ECE Dept Head twice
Bill Davis • Professor at VT for 28 years • Director of Virginia Tech Antenna Group • Past Commission Chair – USNC/URSI Commission A • URSI Meetings Coordinator • Vice Chair 2005 IEEE APS/URSI Sym S ymposium posium
Wireless versus Wireline • Fields of Application – Communications – Sensing and Imaging • Ac Activ tive e – Rad Radar ar • Passi Passive ve – Radio Radiometry metry
– Industrial • Control Ex.: garage door opener • Medical • Heating, cooking, drying, ...
• Communications – Antennas must be used used for: for: • Mobile communications • Very long distances – Space – Remote terrestrial locations
– Antennas are preferred for • • • •
Broadcast and point-to-multipoint comm. Long distance communications Thin routes Portable and personal communications
– The physics of wireless vs. wireline • Loss in wireline is (e- r)2 • Loss in wireless is 1/r2
References W. Stutzman and G. Thiele, “Antenna “Antenna Theory Theory and and Design,” Design,” 2e, Wiley, 1998. T. Rappaport, “Wireless Communications: Principles and Practice,” Prentice-Hall, 1996. K. Siwiak, “Radiowave “Radiowave Propagation and and Antennas for Personal Personal Communic Comm unication ations,” s,” Sec. Ed., Artech Artech Hous House, e, 1998. 1998.
The History of Communications Pre-modern civilization Optical communications: Smoke signals, flags, .... Acoustical communications: Drums [note – all are forms of of digital communicatio communications] ns] 1844 Telegraph (Morse) 1864 Maxwell’s Maxwell ’s equations equations – princip principles les of radio waves 1876 Telephone (Bell) 1887 First antenna (Hertz) 1897 First radio systems (Marconi, Popov)
1901 First transatlantic radio (Marconi) World War II Development of radar 1960 Fiber optics 1980s Wireless reinvented Wireless Wireless
Radio
1900
2000
The Spectrum • Wavelength
(m) = 300 / f(MHz)
(cm) = 30 / f(GHz)
• Frequency bands Band
Frequencies
Wavelength s
VLF
3 KHz
30 kHz
100 km 10 km
LF
30 kHz
300 kHz
10 km
1 km
MF
300 kHz
3 MHz
1 km
100 m
HF
3 MHz
30 MHz
100 m
10 m
VHF
30 MHz
300 MHz
10 m
1m
UHF
300 MHz
3 GHz
1m
10 cm
SHF
3 GHz
30 GHz
1 cm
1 mm
Antenna Performance Parameters • Radiation pattern F( , ) The angular variation of radiation around an antenna Pattern types: Directive (narrow main beam) Omnidirectional Shaped beam Low side lobe
• Directivity D Ratio of power density in the direction of pattern maximum to the Average power density at the same distance from the antenna; i.e. how much more focused the power is than if isotropically distributed.
• Gain G Directivity reduced by losses on the antenna
• Polarization The figure traced out with time by the instantaneous electric field vector. Types: Linear, circular, elliptical, dual (for diversity and reuse)
• Impedance Input impedance at the antenna terminals
• Bandwidth Range of operating frequencies for which performance parameters are acceptable.
• Scanning Movement of the radiation pattern in angular space Types: electronic, mechanical, hybrid
• Mechanical Size, weight, RCS, aerodynamics
• Cost
2. ANTENNA FUNDAMENTALS • What is an antenna? • Connecting to the antenna • Basic properties – Impedance – Gain – Pattern – Polarization
• Fundamental Limits
What is an antenna? • Collection of metal/material objects – Wire – Plates (reflector, dish) – EM Bandgap Materials …
• Absence of metal – Slots – Waveguide apertures
Transformation from Electronics to Space
Determining the Radiation • Start with the current (assume) – Wire: Approximate sinusoid/traveling wave – Plate: Physical Optics approximation or image current
• Giving for example (dipole):
J (r , ω ) = zˆ I cos( β z )δ ( x)δ ( y ) r
r
To the Fields Current Vector Potential
A(r ) = μ r
r
e
− j β r
J (r ' )e ∫ 4π r r
r
V
r
r
E , H
ˆ• r ' j β r r
dv '
The Fields (far field) • Electric Field
H = r
1 μ
r
ˆ × A) (− jβ r
• Magnetic Field
ˆ × H = − jω ( A − r ˆ Ar ) E = −η r r
r
r
Far-Field Conditions • Wavelength
r >> λ • Distance
r >> max(r ' ) = D • Size
r >
2 D 2 λ
⇒ phase <
λ 16
Far Field Observation
Distance Antenna
Projection ~ replace distance
Diameter D
Example – Short Dipole J ~ zˆ I Δ zδ (r ) r
• Current
r
r
e − j β r
• Vector Potential
A = μ
• Magnetic Field
H = j β
• Electric Field
E = j β
• Poynting Vector
S = E × H 2
r
r
r
1
r
4π r
e − j β r 4π r
e − j β r 4π r r
*
zˆ I Δ z
I Δ z sin θ φ ˆ
η I Δ z sin θ θ ˆ
1
= r ˆηβ
2
2
P = ηβ
2
I
2
I
2
Δ z 2 2 2
16π r
Δ z 2
12π
sin 2 θ
Radiation Resistance – Short Dipole • EQUATE – Input Power & Radiation Power
Rrad
=
P rad 1 2 I 2
⇒
ηπ
Δ z
3
λ
2
≈ 40π
2
Δ z
2
λ
Radiation Pattern • Variation of Fields with Elevation (θ) and Azimuth (φ)
F (θ , ϕ ) =
E (θ , ϕ ) max E
= sin θ , for short dipole
θ ,ϕ
• Variations: – Spherical – E-plane & H-plane (or Elevation & Azimuth) – Conical
Patterns
Linear, Principal-Plane Cuts 120
90
60 0.5
150
θ
1 120 30
180
90
1 60 0.5
150
30
0 θ 180
210
330 240
270
300
y = 0 or φ = 0 ° Cu t
120
90
1 60 0.5
150
30
0 φ 180
210
330 240
270
300
x = 0 or φ = 90 ° Cut
0
210
330 240
270
300
z = 0 or θ = 90 ° Cut
Half-Power Beamwidth
Efficiency e=
• Loss
Power Radiated Power Input
– Radiation Resistance
• Mismatch
• Total
q = 1−
Rloss
2
Γ =
=
=
Rrad Rrad + Rloss
1 I
2
∫ J r
S
2
R surface ds
S
4 Z o R Ant
Z Ant + Z o
2
, Z o Real
Efficiency = qe
Directivity & Directive Gain D(θ , φ ) =
F 2 2
Average( F )
=
4π F 2
∫
F 2 (θ , φ )d Ω
θ ,φ
D = max[ D(θ , φ )] D =
4π
Ω A
Ω A = ∫ F 2 d Ω = Beam Solid Angle θ ,φ
Directivity d
= sin θ dθ d d θ
θ
Figure 1-17 Element of solid angle d .
sin θ d
U m
U m
Figure 1-18
A
Antenna beam solid angle
(a) Actual pattern
(b)
Open Circuit Voltage & Effective Length • Open Circuit Voltage r
V OC = − h
*
• E r
• Effective Length r
ˆ × r ˆ× h (θ , φ ) = − r
1
J ( r ' )e ∫ I r
r
ˆ• r ' j β r
Ant
r
=−
E rad jωμ (e − j β r / 4π r ) I
r
dv '
A.
Polarization • Linear, Circular, Elliptical
ˆ E 2 cos(ω t + δ ) E = xˆ E 1 cos ω t + y
• Polarization Factor
r
2
2
r
p = h E / h • E r
r
2
ˆ E x + ˆy Ey = x ˆ E1ejwt + y ˆ E2 ej( t + E=x
E 2
)
2
E 1
Right-handed
Left-handed
Counterclockwise
Clockwise
1
1 2
0
-180°
-135°
-90°
-45°
0°
+45°
+90°
+135°
+180°
Figure 2-37 Polarization ellipses as a function of the ratio E2 /E1 and phase angle with wave approaching. Clockwise rotation of the resultant E corresponds to left-handed polarization (IEEE definition) while counterclockwise corresponds to right-handed polarization.
Gain & Realized Gain
G = eD
• Gain
• Realized Gain
G R
= qeD
• Partial Realized Gain
g R
= pG R = pqeD
A Communication Link
• Friis’ Trans. Formula
P Rcv
=
2
P Xmit GT GR λ (4π R)
2
Effective Area 2
A R (θ , φ ) =
λ
4π
G R (θ , φ )
To Give
P Rcv
=
P Xmit GT AR 4π R
2
Transient Link μ ⎞ ⎛ R ⎞ ∂i (t ) ⎛ vOC = hr (t ) •⎜ ⎟ht ⎜ t − ⎟ ∗ ⎝ 4π R ⎠ ⎝ c ⎠ ∂t r
∗
r
Connections • Connectors (coax, twin-lead) • Balanced vs Unbalanced – Balun
• Feed Network (Arrays) – Phased – True Time-Delay
• Filtering & Impedance Transformation – Circuit & loading – Tapering
Properties • Impedance – Treat as a circuit element – By Reciprocity:
Z Rcv
= Z Xmit
– Induced EMF Z
Z =
1 2
I
∫ J • E ( J )d Ω r
Ant
r
r
Properties • Patterns
F Rcv
= F Xmit
Pattern Reciprocity
Fundamental Limits • A bit of Controversy – Chu (1948)
– McLean
+1 Q= 3 3 2 2 β a ( β a + 1) 2
2 β a
2
Q=
1 3
β a
3
+
1 β a
Antenna Fundamental Limit
2
10
Patch Inverted F Dual Inverted F 1
10 Q n o i t a i d a R
Planar Inverted F
Dipole
Goubau Foursquare Wideband, Compact, Planar Inverted F 0
10
-1
10
0
McLean Grimes Chu 0.5
1
1.5
2
2.5
ka
2. ANTENNA ELEMENTS • The Four Antenna Types – Electrically Small Antennas – Resonant Antennas – Broadband Antennas • Frequency Independent • Ultra Wideband
– Aperture Antennas
The Four Antenna Types • Electrically Small Antennas Examples Short dipole
Small loop
Properties Low directivity Low input resistance Low efficiency and gain
• Resonant Antennas Examples Dipole
Microstrip antenna
Properties Low to moderate gain Real input impedance Low efficiency and gain
Yagi
Monopoles and Images
• Broadband Antennas Examples (Frequency Independent) Spiral
Log Periodic Dipole Array
There are two types of broadband antennas, ones that have frequency independent performance and ones that preserve signal properties in the time domain (UWB) Properties Low to moderate gain Real input impedance Low efficiency and gain
• Aperture Antennas Examples Horn antenna
Parabolic reflector antenna
Properties Low to moderate gain Real input impedance Low efficiency and gain
3. ANTENNA ARRAYS A. Array Basics B. Arrays of Isotropic Elements C. Inclusion of Element Effects D. Mutual Coupling E. Phased Arrays
A. Array Basics Def.: Array Antenna. An antenna comprised of a number of identical radiating elements in a regular arrangement and excited to obtain a prescribed radiation pattern. Advantages of arrays: • Many small antenna elements instead of one lar ge mechanical structure
• Scanning at electronic speeds is possible • Multiple user (target) tracking is possible • Many geometries, including conformal, are possible Problems: • A feed network is required with its losses and bandwidth limitations • Mutual coupling between elements affects performance and complicates design • Computer control may be necessary
General array configuration with feed network Phase shifter Attenuator
Feed network
Figure 3-1 A typical linear array. The symbols and indicate variable phase shifters and attenuators. The output currents are summed before entering the receiver
Receiver
B. Linear Arrays of Isotropic Elements • General array configuration of isotropic elements Rays
Phase =
1e
j ξ 0
ξ0
1e
Reference wavefront
ξ1
j ξ1
1e
j ξ n
ξn
Wavefronts
θ
Ph ( I n ) I n
I 0 e
I n e
j ξ0
j ξ1
In e
Figure 3-2 Equivalent configuration of the array in Fig. 3-1 for determining the array factor. The elements of the array are replaced by isotropic point sources.
j ξ n
Array factor
AF
I0 e
j
0
j
I1 e
1
I2 e
j
2
...
(3-3)
ξ n = phases at element n due to incoming wave In
= complex current representing the feed network amplitude and phase at element n
• Two element arrays of isotropic elements of various spacings and phasings 0°
d
d
45°
90°
135°
180°
1 8
1 4 Elements
d
d
3 8
d
I 1
I 1
1 2
From [Kraus] d
5 8
d 1
• General uniformly excited, equally spaced linear array (UE,ESLA)
r s o c
θ
d
θ
d
0
θ
z
d
1
2
Figure 3-7 Equally spaced linear array of isotropic point sources. N 1
AF
I0
I1 e j
d cos
I2 e j
2 d cos
In e j
nd cos
(3-14)
n =0
Now consider the array to be transmitting. If the current has a linear phase progression (i.e., relative phase between adjacent elements is the same), we can separate the phase explicitly as jna
In
where the n +
1th
An e
element leads the
nth
element in phase by a. Then (3-14) becomes
N 1
AF =
(3-15)
An e
jn
d cos
a
(3-16)
n 0
Define Then
d cos + a
(3-17)
N 1
An e jn
AF = n 0
(3-18)
Universal array factor
Properties of the universal array factor • The array factor is periodic in 2 Proof: AF
2
A ne
jn
2
An e jn e jn2
Ane jn
AF • Visible region extent:
180°
θ
0
1 < cos θ < 1 βd < βd cos θ < βd α βd <
< α + βd
• Exactly one period of the array factor appears in the visible region when the
element space is /2. Proof: Width of the visible region = 2 d AF period = 2 For one period visible: 2 = 2 d = 2(2 / )d d = /2 For d > /2, grating lobes may appear in the visible region depending on For d
.
, grating lobes will appear in the visible region
The general array factor expression for a UE,ESLA A0
A1
A2 N 1
AF = A0
e
jn
n 0
A0 e
j N 1
/2
sin N /2 sin
/2
The expression is maximum for
AF
f
=0
=0
A0 1 + 1 +
sin N /2 N sin
/2
+1
UE, ESLA
A0 N
(3-33)
This is the normalized array factor for an N element UE, ESLA that is centered about the coordinate origin.
Universal array factors for N = 3, 5, 10
f
1.0
f
1.0
N = 5 N = 3
( a)
( b)
0 0
2 3
1.0
0 0
2
4
0.4
0.8
1.2
1.6
2
3
f N = 10
Figure 3-11 Array factor of an equally spaced, uniformly excited linear array for a few array numbers. (a) Three elements. (b) Five elements. (c) Ten elements.
(c) 0 0 0.2 0.4 0.6
0.8
1.2 1.4 1.6 1.8
2
Example 3-5 Four-element array steered to 120 ° z 1 (a)
j /2
d
1e
sin 2
f
1.0
(b)
j 3 /2
1 e j
1e
4 sin
2
3
-
2
2
2
2
Figure 3-12 Array factor for a four-element, uniformly excited, equally spaced phased array (Examples 3-5). (a) The array excitations. (b) Universal pattern for N = 4. (c) Polar plot for d = /2 and = /2.
θ θo
(c)
= 120°
d= 2
z
• Scanning the pattern of a linear array in space The main beam maximum occurs for = 0. Let θo be the value of θ in the direction of the beam maximum. Then ψ = 0 = βd cos θo + α
α = βd cos θo
(3-36)
This element-to-element phase shift will scan the main beam peak to θ = θ0. Often, we express
as
ψ = βd (cos θ - cosθo )
(3-38)
• Beamwidth of the main beam For Nd >> λ
(Nd = L = length of the array)
HP
0.886(λ/Nd) csc θo
HP
2 0.886 λ /Nd
1/2
near broadside
(3-45)
at endfire
(3-46)
Example N = 20 d = λ /2 0.886 λ/Nd
0.886 λ/ 20λ /2
0.0886 radians
θo
90
HP = 0.0886 r = 0.0886 180/
θo
0
HP = 2 0.0886
1/2
= 0.595 180/
5.1° 34.1°
• Directivity of Uniformly Excited, Equally Spaced Linear Arrays 20
sin N /2 D
4
1 N
N 1
2 N
2 m 1
D
/2
N sin
4
N - m
sin m β d cos m mβd
(3-78) λ n , 2 Then D N
If
=0
d
L 2 for N 10 λ
α= 0
15
=0 )
D ( y 10 t i v i t c e r i D
N = 10 9 8 7 6 5 4
5
3
Figure 3-20 Directivity as a function of element spacing for a broadside array of isotropic elements for several element numbers N.
0 0
0.5
1.0
1.5
d λ
The directivity of a broadside array of isotropic elements is approximated by
D
2
L
λ
2
Nd
λ
broadside
(3-80)
where L = Nd is the array length. This is a straight-line approximation to the curves
2.0
• Nonuniformly excited, ESLA N-1
An e jn
AF
βd cos
n=0 An
1.0
Uniform
0.5
(a)
0
1: 1 : 1 : 1 : 1
z
0
1/2
3/2
2
An
1.0
Triangular (b)
z
0
1/2
3/2
2
An
1.0 0.5
(c) 0
1/2
3/2
2
1/2
3/2
2
z
An
1.0 0.5
(d)
0
z
0
1.0
0 0
1/2
D HP
3/2
2
z
Dolph-Chebyshev for a side lobe level of -30dB 1 : 2.41 : 3.14 : 2.41 : 1
D
5
0.75 0.25 0.50
4.26
HP
20.5 SLL
180°
Figure 3-24 Current distributions Binomial corresponding to the patterns of Fig. 3-23. The current phases are 1:4:6:4:1 zero ( = 0). Currents are normalized to unity at the array center. (a) Uniform. (b) Triangular. Dolph-Chebyshev (c) Binomial. (d ) Dolphfor a side lobe level of -20dB Chebyshev (SLL = -20 dB). 1 : 1.61 : 1.94 : 1.61 : 1 (e) Dolph-Chebyshev.
An
0.5
(e)
1:2:3:2:1
26.0
1.00
(a) Uniform currents, 1 : 1 : 1 : 1 : 1.
180°
0.25
0.50
0.75
3.66
HP
SLL
12 dB
0°
D 19.1 dB
30.3 SLL
1.00
(b) Triangular current amplitude
distribution, 1 : 2 : 3 : 2 : 1.
0°
180°
0.25
0.50
0.75
dB
1.00 0°
(c) Binomial current amplitude
distribution, 1 : 4 : 6 : 4 : 1.
Figure 3-23 Patterns of several uniform phase (θo = 90°), equally spaced (d = /2) linear arrays with various amplitude distributions. The currents are plotted in Fig. 3-24.
D D HP
4.68 SLL
0.25
180°
0.50
HP
23.6
0.75
4.22 26.4 SLL
30 dB
-20 dB
1.00 1.00
0°
0.50 180°
0.75
0.25
(d) Dolph-Chebyshev
(e) Dolph-Chebyshev
Current amplitude distribution, 1 : 1.61 : 1.94 : 1.61 : 1 for a side lobe level of -20 dB.
Current amplitude distribution, 1 : 2.41 : 3 .14 : 2.41 : 1, with a side lobe level of -30 dB.
Fig. 3-23 (continued)
We can draw a general conclusion from the foregoing examples that applies to antennas in general: As amplitude taper increases: Beamwidth increases Directivity decreases (as a consequence of beamwidth increasing) Sidelobes decrease Current envelope:
Pattern:
0°
Directivity of nonuniformly excited, isotropic element arrays General case of unequally spacings and nonuniform phasings 2
N 1
Ak
(3-91)
k =0
D N 1 N 1
j
Am Ap e
m
sin
p
zm
zp
zm
m =0 p =0
z p
Equal spacings and broadside operation 2
N 1
Ak D
k =0 N 1 N 1
Am Ap m =0 p =0
sin
m m
p p
d
n
0,
zn
nd
(3-92)
d
Spacings a multiple of a half-wavelength and broadside N 1
2
An D
n =0
d
N 1
An
2
2
,
(3-93)
,...
n =0
C. Inclusion of Element Effects Principle of Pattern Multiplication: The pattern of an array (array pattern, F) consisting of similar elements is the product of the pattern of one of the elements (the element pattern, ga) and the pattern f the array of isotropic elements with the same locations, relative amplitudes and phases as the original array (the array factor, f). Pattern F
Element pattern ga
Array factor f
F(θ, ) = ga (θ, ) f(θ, )
(3-36)
Collinear elements Parallel elements Example: Two collinear short dipoles
| Io
λ 2 1
|
θ
x z
I 1 1 (a) The array.
Element pattern
Array factor
Total pattern z
θ
y
Figure 3-18 A linear array of parallel line sources. sin
sin
cos
2
cos
cos
2
cos
(b) The pattern. Figure 3-17 Array of two half-wavelength spaced, equal amplitude, equal phase, collinear short dipoles (Example 3-8).
Parallel half-wave dipoles For a half-wave dipole along z-axis z
cos
/2 cos
half
sin
(2-8)
wave dipole
For an array of parallel half-wave dipoles it is best to orient the elements along the x-axis. Then the element pattern is expressed as x
cos
g a
z
(3-68)
sin sin
cos ga
/2 cos
,
cos
cos
sin = 1 - sin /2 sin
1- sin 2
2
cos
2
cos
cos 2
(3-69)
Prob. 3.3-2 2 λ
1
cos From (2-7)
ga θ
From (3-13) f θ
cosθ
2
sinθ
cos
cosθ cos
So F θ
1
g a θ f θ
cos θ 2 cos sinθ
cos
x
ga
f
F
=
z
z
z
(Fig. 2-5b)
(Fig. 3-6c)
• Method used in practice to produce a single endfire beam x
y
z
λ 4 Ground plane
Image theory solution
The array factor is that of Example 3-2 y
x
yz-plane (array
factor) -1
1
d=
λ 2
-1
1
d=
λ 2
Problem 3.7-7 uses array theory to find the patterns
xz-plane (including
/2 dipole effect)
• Directivity of arrays of real elements There is no general exact formula for the directivity of arrays including the element pattern. The following often- quoted approximate directivity formula must be used with caution
D De Di De
directivity of a single element
Di
directivity of the array with isotropic elements
Example: 4-element, broadside array of collinear short dipoles with spacing d = /2 (3-83) gives D = 5.6, an exact answer
D DeDi
1.5
4
6
• Base Station Collinear Arrays
d
0.72
From code L
D = 7.75 dBi 2
d = 0.72
= 5.6dBd
Approximate directivity D
DeDi = 9.5 d B
1.64(5.4) 7.3 dBd
The tower enhances the gain by making pattern more directive
8.9, u sing Fig. 3-20
Base station with elements in opposing pairs and clocked 90 degrees around the tower to produce nearly an omnidirectional pattern
D. Mutual Coupling
1
2
m
N
Feed network
Figure 3-26 (a) Mechanisms for coupling between elements of an arra y. There are three mechanisms r esponsible for mutual coupling: Direct free space coupling between elements Indirect coupling due to scattering by nearby objects Coupling through the feed network In many arrays the elements are impedance matched to the feed network and feed coupling can be ignored. Then the array can be modeled with independent generators, leading to the conventional circuit N port representation. V1 Z11 I1 Z12 I2 Z1N IN
V2
Z12 I1
Z 22 I2
Z2N IN
VN
Z 1N I1 Z1N I2
Z NN IN
where the mutual impedance is Vm with I i 0 for all i except i = n Zmn Im and reciprocity has been assumed through Z mn = Znm
The input impedance of the mth element in an array with all elements active and mutual coupling included is
V m Vm
Zm
Im
Z m1
I1 Im
Zm2
I2 Im
ZmN
1N
Z g m
(3-103)
1m
This is often called active impedance or driving point impedance.
Z m
I m
V m g
Note that active impedance depends on mutual impedances between elements as well as the excitations of all elements. This dependence includes the current phases and thus scan angle in phased arrays. The effects of mutual coupling include: •
The impedance of an element in an array differs from its free space value and depends on that array scan angle (element phases) and the element location. •
•
The pattern of an element is changed from its isolated pattern and depends on array position.
Polarization characteristics deteriorate.
Example: Mutual coupling effect on a 12-element, half-wave spaced linear array Current generator Voltage generator (loaded)
y
0 -10 dB
x
-20
z
x
= 180°
Figure 10-27 Linear array pattern with main beam steered to o = 45° and ideal current generators (solid curve) compared to patterns from an array with voltage generators for 72- loaded voltage generator excitations (dashed cur ve). Also see: D. Kelley and W. Stutzman, “Array antenna pattern modeling methods that include mutual coupling effects,” IEEE Trans. On Ant. And Prop., Dec. 1993.
= 0°
E. Phased Arrays 90°
90° θo=
θo=
90°
1.00 0.75 0.50 0.25 0°
180°
(a) θo= 90° 90°
(c) θo= 75°
(b) θo= 90° 90° θo=
180°
1.00 0.75 0.50 0.25 0°
180°
z
θo=
30°
0.75 1.0 0.50 0.25 0°
75°
0°
1.0 0.50 0.75 0.25 0°
180°
z
(f) Endfire ( θo = 90°) (d) θo = 30° (bifurcated pattern)
(e) Endfire ( θo = 0°)
Figure 3-32 Example of phase-scanned patterns for a five-element linear array along the z-axis with elements equally spaced at d = 0.4 and with uniform current magnitudes for various main beam pointing angles θo.
Important point to remember: The array factor scans inside the envelope o f the fixed element pattern Example: Linear array of four broadside elements spaced 0.7 apart. The element pattern is g(θ) = (cos θ)2. Broadside operation
Fmag(θ)
Scanned 30° off broadside
g(θ) 1
1
0.8
0.8
0.6
0.6
0.4
0.4
0.2
0.2
0 -90-80 -70 -60-50 -40-30-20 -10 0 10 20 30 40 50 60 70 80 90 θ
0 -90-80-70 -60-50-40-30-20 -10 0 10 20 30 40 50 60 70 80 90 θ
• Basic Feed Types
(b) Series feed.
(a) Parallel, or corporate, feed.
Primary transmitting antenna Pickup antenna (c) Space feed.
Secondary transmitting antenna
(d) Parallel-series feed.
Figure 3-33 Types of array feed networks.
Phased array example: AWACS (Airborne Warning and Control System) Mounted on top of aircraft such as E-3A, E-2C for aerial surveillance and detection of bombers and low flying fighters coming in over north pole First operational radar antenna with very low sidelobes: -40 dB Slotted waveguide array with 4000 slots Scanned in vertical plane using ferrite phase shifters Rotated for azimuth coverage Next generation: phased array conforming to the aircraft
E-2C
• Beam switched scanning
Go 4
2
3
Pe 1 x 4 Switch
1 (a)
Go
Pe
Pe
1
Ps ERP = PTGT = GePs Ps
Go
4
Rotman Lens or Butler Matrix
10 Log N = 6 dB for N
4
2
ERP = PT GT
NGo NPo
3
4
1 x 4 Switch (b)
Ps
Pe
Ps 4
4GsPs
Ps
(c)
Ps
Switched antenna system vs linear array configurations: (a) switched antennas; (b) multiple-beam array; (c) steered-beam array. [Microwave Journal , January 1987]
• Digital Beam Forming
Receiving Antenna
Array
RX
ADC
RX
ADC
Beams
Digital Beamformer
RX
ADC
Weights Controller Calibration Tone Directions of Look
• Scanning arrays of wideband elements [Stutzman and Buxton, Microwave Journal, Feb. 2000.]
1 ) λ
/ d ( g n i c a p S t n e m e l E d e z i l a m r o N
0
l s i r a p S
0.9
6.4
0.8
14.5 ..
s o u u n S i
0.7 0.6
25.4 41.8
r e u a q u r s F o
0.5
90
0.4 0.3
e l g n A n a c S m u m i x a M
0.2 0.1
0
1
1.5
2
2.5
3
Bandwidth (f/f ) L
The future for phased arrays - Wideband, multifunctional arrays will be used - Intelligence will be integrated with arrays, creating “smart arrays” that can adapt to changing conditions and faults - Radiating elements will be printed, giving low cost and uni form geometrical construction (see figure) - Feed networks will make use of integrated fabrication techniques such as MMIC. - Elements and feeds will be integrated together - Beam steering: Low cost phase shifters: MEMS, Ferroelectric, . . . Photonic feeds Digital beam forming will be very popular as RF and DSP module performance and cost improve
2x2 array of Foursquare elements
Fourpoint element
capable of 2:1 bandwidth and dual polarization
5. SYSTEM CONSIDERATIONS • Friis Transmission Equation • Propagation in real links • Factors in selecting an operating frequency
• Factors in operating frequency selection – Propagation and link budget considerations • VHF and below for long distance, narrow bandwidth • UHF and above for wide bandwidth • Above 10 GHz, atmospheric losses are high
– Antenna considerations • Very long distance point-to-point communications require high frequencies to enable large antenna gains
– Regulatory issues • Must use allocated bands • Licensed vs. unlicensed bands
6. WIRELESS APPLICATIONS • Base Station Antennas for Land Mobile • Antennas for Satellite Communications • Vehicle Antennas • Antennas for Personal Communications • UWB – Some Concepts
• Radiation Safety
Base Station Antennas for Land Mobile Radio • Cell Coverage Types
Omnidirectional
Smart antenna that forms beams on users
Sectorized
• Omnidirectional pattern base station antennas – Pattern is constant in azimuth and narrow in elevation – Usually realized with collinear array of dipoles Example: Cellular base station
Antel Cellular Base Station Antenna 870-970 MHz Collinear dipoles 136 in long 10 dBd gain 1.25 deg downtilt
• Sector base station antennas Typical 120 deg sectors, VP
Polarization diversity 120 deg sectors, dual slant 45 deg LP
• Sector (“panel”) antennas – Elements • • • •
Dipoles in front of a ground plane Log periodic dipole (LPDA) and LPDA vees Patches Fourpoint antenna (VT) covers both cellular and PCS with dual polarization
– Considerations • Bandwidth (VSWR<2) • Power handling • Intermodulation products
DB Products LPDA vee antenna
• A new wideband base station antenna developed at VTAG: – Minimum number of antennas on tower – Dual-polarized for diversity – Multi-functional capability (800 ~ 2200 MHz) – Low profile and compact – Printed antenna on a PCB
Front view
Tuning plate on back
Fourpoint Antenna Data
AMPS, GSM, DCS, PCS, etc
Measured Measured Radiation Radiation Patterns Patterns H-plane
E-plane
900 MHz • HPBW: 60 ~ 80
1800 MHz
• Low cross-pol, less than -30 dB
• Satellite communications antennas Satellite services FSS – Fixed Satellite Service [gateways, VSATs] BSS – Broadcast Satellite Service (DBS, DTH) Mobile (Iridium, Orbcom, Inmarsat)
Gain of spacecraft antenna Global coverage US coverage
19 dB 34 dB
Ground station antennas Reflectors are used above a few GHz Small offset reflectors for VSAT Large, dual reflectors for gateways
Direct Broadcast system from DirecTV manual ________________________ 12.2 -12.7 GHz downlink to user 17.3-17.8 GHz uplink from gateway Dual circularly polarized
Vehicular Antennas • Broadcast reception Traditional 31” (0.003 at AM) fender mount whip antenna is vanishing. New cars have mostly on-glass antennas
Example of a Ford rear window glass antenna
• Two-way land mobile vehicle antennas VHF and below Short monopole Quarter-wave monopole
Example quarter-wave monopole
UHF Quarter-wave monopole 5/8 over ¼ wavelength
Example 5/8 over ¼ wavelength
• Aircraft antennas Example: Commercial MD-80 airplane
• Antennas for fixed wireless – Access points and terminals Omnidirectional antennas Collinears for access points Stubby or planar antennas for terminals High gain (20 dB or more) – reflectors Moderate gain Yagi Stub loaded helix antenna 75% volume reduction Circularly polarized
Example; Stub loaded helix antenna www.frc-corp.com
• Antennas for personal communications VHF Short monopole Loops, small and/or loaded Normal mode helix UHF Monopole Normal mode helix Patch Inverted-L Planar inverted-L Embedded
Perhaps the most popular antenna for cell phones is the stubby antenna with an extendable wire antenna
Nokia Patent 5,612,704
• Inverted-L family of antennas
Inverted-L
Inverted-F
Dual Inverted-F
Planar Inverted-F
Wideband Compact PIFA
VT patent 6,795,028
Antenna
Bandwidth (%)
Patch
1
IFA
2
DIFA
4
PIFA
8
Dipole
12
WC PIFA
43
UWB Antennas
TEM Horn
Tapered Slot (Vivaldi)
Ridged Horn
BiCone
Impulse Radiating Antenna (IRA)
Half-Disk
PICA & VSWR 5 Simulation Measurement
4.5 4
3.5
3 2.5
2 1.5
1
2
4
6
8
10
12
14
16
18
20
Typical Responses – TEM Horn Reflection
Transmission Link Frequency Response s 21
Return Loss 0
0
) B-10 d ( e d -20 u t i l p m-30 A
-40 0
) B d (
1 2
-50
s
2
4
6
8 10 12 Frequency (GHz)
14
16
18
20
-100
Phase - Delay Phase
2
4
6
2
) s e e 0 r g e d ( e s -200 a h P
8 10 12 Frequency (GHz) Impulse Response s
9
200
-400
0
x 10
14
16
18
20
14
16
18
20
21
)
1 -
s 1 ( e d u t i l p 0 m A
0
2
4
6
8 10 12 Frequency (GHz)
Phase
14
16
18
20
-1
0
2
4
6
8
10 12 Time (nsec)
Transient Transmission
Radiation Safety & Interference • SAR – Specific Absorption Rate – Power per area absorption • HAC – Hearing Aid Compatibility – Buzzing and related noise in Hearing Aid • System Interference and Cross-Modulation
Facilities Measured Near Field Pattern
Calculated Far Field Pattern
INSTRUMENTATION •ANTCOM 7+1 axis near field –far field scanner •Agilent 8510, 8511, 8530 Network Analyzer
