Sunil Thesis

INTERNATIONAL SOCIETY OF THESIS PUBLICATION A Society of Research Publication

D E S I G N & A N A LY S I S O F M I C R O S T R I P PAT C H A N T E N N A U S I N G M E TA M A T E R I A L SUNIL KUMAR THAKUR AMIE(EC),ME(CCN)

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DESIGN & ANALYSIS OF MICROSTRIP PATCH ANTENNA USING METAMATERIAL

ABSTRACT This thesis work focuses mostly on design and analysis of microstrip patch antenna using metamaterial as well as effects of slots made on patch antenna to improve the banwidth. Microstrip patch antenna are preferred over other antennas in todays modern word scenario for their compatibility to be fit in Mobile , Aircraft , Satellites owing to very small sizes. Hence design and development of superior and cost effective microstrip patch antenna has become an active research area. We present characteristics of microstrip patch antennas on metamaterial substrates loaded with complementary split-ring resonators (CSRRs). The proposed antenna utilizes CSRRs in the ground plane altering the effective medium parameters of the substrate. Simulation results were verified by experimental results. The experimental results confirm that the CSRR loaded patch antenna achieves size reduction and keeping the bandwidth intect as well. The radiation properties of a circular patch antenna with U-slot designed on glass epoxy FR-4 substrate are obtained and compared with that of a normal circular patch antenna designed under identical conditions. The modified antenna not only resonates at two different frequencies but also presents marked Improvement in the bandwidth. The, return loss and VSWR are measured for simple circular patch antenna and circular patch antenna with U-shaped slot. Reasonably good matching of modified antenna with the feed network is obtained for both the resonance frequencies. The designed antennas have been characterized using the commercially available software IE3D. With the help of this we discuss reflection coefficient, transmission coefficient, VSWR, radiation pattern etc. All the designs have the same dielectric constant ( ) 4.4, substrate thickness of 1.6 mm and loss tangent of 0.02.

pg. 2


CONTENTS Nomenclature List of Figures Chapter 1

Introduction of microstrip patch antenna and metamaterial 1.1 Microstrip patch antenna 1.1.1 Introduction 1.1.2 Advantage and Disadvantage 1.1.3 Basic principle of operation 1.1.4 Resonant Frequency 1.1.5 Feed technique 1.1.5 (a) Microstripline Feed 1.1.5 (b) Coaxial feed 1.1.5 (c) Aperture couple Feed 1.1.5 (d) Proximity couple Feed 1.1.6 Method of analysis 1.1.6 (a) Transmission line method 1.2 Metamaterial 1.2.1 Negative effective refractive index 1.2.2 Design strategies for NRM 1.2.2 (a) Thin metallic wire 1.2.2 (b) Swiss roll structure 1.2.2 (c) Split ring resonators 1.2.2 (d) Complementary SRR 1.3 Organization of the thesis

Chapter 2 Design and analysis of Microstrip Patch Antenna 2.1 Design procedure 2.1.1 Summary of design parameters 2.1.2 Concluding remarks Chapter 3 Microstrip patch antenna loaded with Complementary SRR 3.1 Introduction 3.1.1 Proposed antenna configuration 3.1.2 Design and analysis of complementary pg. 3

Page No 5 -6 07 07 07 08 09 10 10 10 11 11 12 12 12 15 15 16 16 17 18 19 19

20 20 21 23

24 24 24


SRR loaded patch antenna 3.1.3 Concluding remarks Chapter 4 Compact dual frequency wide band circular Patch antenna with U slot 4.1 Introduction 4.1.1 Proposed antenna geometry 4.1.2 Design of circular patch antenna 4.1.3 Design of circular patch antenna With U slot 4.1.4 Concluding remarks Chapter 5 Conclusion and suggestion for future works 5.1 Concluding remarks 5.2 Suggestion for future works References

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25 29

30 30 30 31 32 34 35 35 35 36


LIST OF FIGURES Figure: 1.1

Structure of microstrip patch antenna

Figure: 1.2

Common shapes of microstrip patch antenna

Figure: 1.3

Microstrip line feed

Figure: 1.4

Probe fed rectangular microstrip patch antenna

Figure: 1.5

Aperture coupled feed

Figure: 1.6

Proximity feed

Figure: 1.7

Microstrip line

Figure: 1.8

Electric field lines

Figure: 1.9

Microstrip patch antenna

Figure: 1.10(a)

Top view(patch antenna)

Figure: 1.10(b)

Bottom view(patch antenna)

Figure: 1.11

Metallic wire meshes with negative dielectric permitivity

Figure: 1.12

Swiss roll structure

Figure: 1.13(a)

Circular structure

Figure: 1.13(b)

Square structure

Figure: 1.14

Frequency dependence of effective permittivity for a SRR

Figure: 1.15(a)

Circular structure

Figure: 1.15(b)

Square structure

Figure: 2.1

Simulated geometry of patch antenna

Figure: 2.2

Photograph of patch antenna(top layer)

Figure: 2.3

Photograph of patch antenna(Bottom layer)

Figure: 2.4

Current distribution

Figure: 2.5

Return loss Vs frequency of patch antenna

Figure: 2.6

VSWR v/s frequency of patch antenna

Figure: 2.7

Comparative graph of return loss Vs frequency

Figure: 3.1

Configuration of metamaterial substrate antenna

Figure: 3.2

Comparision of simulated return loss

Figure: 3.3

Simulated geometry of patch antenna

Figure: 3.4

Unit cell CSRR

pg. 5


Figure: 3.5 (a)

Simulated geometry of patch loaded with CSRR (top layer)

Figure: 3.5 (b)

Simulated geometry of patch loaded with CSRR (bottom layer)

Figure: 3.6 (a)

photograph of fabricated antenna (upper layer)

Figure: 3.6 (b)

photograph of fabricated antenna (upper layer)

Figure: 3.7 (a)

photograph of fabricated antenna (bottom layer)

Figure: 3.7 (b)

photograph of fabricated antenna (bottom layer)

Figure: 3.8

Return loss Vs frequency of a normal patch antenna

Figure: 3.9

Return loss Vs frequency of a CSRR loaded patch antenna

Figure:3.10

Phase change at1.4GHz of a CSRR loaded patch antenna

Figure: 3.11


Figure: 4.1

Proposed geometry of circular patch antenna with U slot

Figure: 4.2

Simple circular patch antenna

Figure: 4.3

Variation of S11 with frequency of a circular patch antenna

Figure: 4.4

Circular patch antennas with U slot made at the patch

Figure: 4.5

Photograph of U slot circular patch antenna (top layer)

Figure: 4.6

Photograph of U slot circular patch antenna (bottom layer)

Figure: 4.7

Variation of S11 with frequency of a U slot circular patch antenna

Figure: 4.8


Figure: 4.9

Variation of VSWR with frequency

pg. 6


CHAPTER ONE INTRODUCTION OF MICROSTRIP PATCH ANTENNA & METAMATERIAL

1.1 MICROSTRIP PATCH ANTENNA Microstrip antennas are attractive due to their light weight, conformability and low cost [7]. These antennas can be integrated with printed strip-line feed networks and active devices. This is relatively new area of antenna engineering. The radiation properties of micro strip structures have been known since the mid 1950‟s.[1] The application of this type of antennas started in early 1970‟s when conformal antennas were required for missiles. Rectangular and circular micro strip resonant patches have been used extensively in a variety of array configurations. A major contributing factor for recent advances of microstrip antennas is the current revolution in electronic circuit miniaturization brought about by developments in large scale integration. As conventional antennas are often bulky and costly part of an electronic system, micro strip antennas based on photolithographic technology are seen as an engineering breakthrough. 1.1.1 INTRODUCTION In its most fundamental form, a microstrip Patch antenna consists of a radiating patch on one side of a dielectric substrate which has a ground plane on the other side as shown in Figure 1.1. The patch is generally made of conducting material such as copper or gold and can take any possible shape. The radiating patch and the feed lines are usually photo etched on the dielectric substrate. Patch

Dielectric of height h

Ground plane

Figure: 1.1 structure of a microstrip patch antenna

In order to simplify analysis and performance prediction, the patch is generally square, rectangular, circular, triangular, and elliptical or some other common shape as shown in Figure 2.2.For a rectangular patch, the length L of the patch is usually 0.3333λo< L < 0.5 λo, where λo is the free-space wavelength. The patch is selected to be very thin such that t << λo (where t is the patch thickness). The height h of the dielectric substrate is usually 0.003 λo≤h≤0.05 λo.

pg. 7


The dielectric constant of the substrate (εr) is typically in the range 2.2 ≤ εr ≤ 12. microstrip patch antennas radiate primarily because of the fringing fields between the patch edge and the ground plane. For good antenna performance, a thick dielectric substrate having a low dielectric constant is desirable since this provides better efficiency, larger bandwidth and better radiation. However, such a configuration leads to a larger antenna size. In order to design a compact microstrip patch antenna, substrates with higher dielectric constants must be used which are less efficient and result in narrower bandwidth. Hence a trade-off must be realized between the antenna dimensions and antenna performance.

Figure: 1.2 Common shapes of microstrip patch antenna [7]

1.1.2 ADVANTAGE AND DISADVANTAGE Microstrip patch antennas are increasing in popularity for use in wireless applications due to their low-profile structure. Therefore they are extremely compatible for embedded antennas in handheld wireless devices such as cellular phones, pagers etc... The telemetry and communication antennas on missiles need to be thin and conformal and are often in the form of Microstrip patch antennas. Another area where they have been used successfully is in Satellite communication. Some of their principal advantages are given below: [7] • Light weight and low volume. • Low profile planar configuration • Low fabrication cost, • Supports both, linear as well as circular polarization. • Can be easily integrated with microwave integrated circuits • Capable of dual and triple frequency operations. • Mechanically robust when mounted on rigid surfaces. Microstrip patch antennas suffer from more drawbacks as compared to conventional antennas. Some of their major disadvantages discussed by Garg [9] are given below: • Narrow bandwidth pg. 8


• Low efficiency • Low Gain • Extraneous radiation from feeds and junctions • Poor end fire radiator except tapered slot antennas • Low power handling capacity. • Surface wave excitation Microstrip patch antennas have a very high antenna quality factor (Q). It represents the losses associated with the antenna where a large Q leads to narrow bandwidth and low efficiency. Q can be reduced by increasing the thickness of the dielectric substrate. But as the thickness increases, an increasing fraction of the total power delivered by the source goes into a surface wave. This surface wave contribution can be counted as an unwanted power loss since it is ultimately scattered at the dielectric bends and causes degradation of the antenna characteristics. Other problems such as lower gain and lower power handling capacity can be overcome by using array configuration forthe elements. 1.1.3 BASIC PRINCIPLES OF OPERATION The metallic patch essentially creates a resonant cavity, where the patch is the top of the cavity, the ground plane is the bottom of the cavity, and the edges of the patch form the sides of the cavity. The edges of the patch act approximately as an open-circuit boundary condition. Hence, the patch acts approximately as a cavity with perfect electric conductor on the top and bottom surfaces ,and a perfect “magnetic conductor” on the sides. This point of view is very useful in analyzing the patch antenna, as well as in understanding its behavior. Inside the patch cavity the electric field the electric field is essentially z directed and independent of the z coordinate. Hence, the patch cavity modes are described by a double index (m, n). For the (m, n) cavity mode of the rectangular patch the electric field has the form Ez(x,y) = Amn Cos (

) Cos (

)

-------------------------------- (i)

Where L is the patch length and W is the patch width. The patch is usually operated in the(1, 0) mode, so that L is the resonant dimension, and the field is essentially constant in the y direction. The surface current on the bottom of the metal patch is then x directed, and is given by For this mode the patch may be regarded as a wide microstrip line of width W, having a resonant length L that is approximately one-half wavelength in the dielectric. The current is maximum at the centre of the patch, x = L/2, while the electric field is maximum at the two“ radiating” edges, x = 0 and x = L. The width W is usually chosen to be larger than the length (W =1.5 L is typical) to maximize the bandwidth, since the bandwidth is proportional to the width. (The width should be kept less than twice the length, however, to avoid excitation of the (0,2) mode.) At first glance, it might appear that the microstrip antenna will not be an effective radiator when the substrate is electrically thin, since the patch current in will be effectively shorted by the close proximity to the ground plane. If the modal amplitude A10 were constant, the strength of the radiated field would in fact be proportional to h. However, the Q of the cavity increases as h decreases (the radiation Q is inversely proportional to h). Hence, the amplitude A10 of the modal field at resonance is inversely proportional to h. Hence, the strength of the radiated field from a resonant patch is essentially independent of h, if losses are ignored. The resonant input resistance will likewise be nearly independent of h. This explains why a patch antenna can be an effective radiator even for very thin substrates, although the bandwidth will be small.

pg. 9


1.1.4 RESONANT FREQUENCY The resonance frequency for the (1, 0) mode is given by

fo =

------------------ ------- 1.1.4 (a)

Where c is the speed of light in vacuum. To account for the fringing of the cavity fields at the edges of the patch, the length, the effective length Le is chosen as

Le= L + 2ΔL------------------ ------- 1.1.4 (b) The formula for the fringing extension [6] is --------------- 1.1.4 (c) where

----------1.1.4(d) 1.1.5. FEED TECHNIQUES Microstrip patch antennas can be fed by a variety of methods. These methods can be classified into two categories:CONTACTING AND NON-CONTACTING. In the contacting method, the RF power is fed directly to the radiating patch using a connecting element such as a microstrip line. In the non-contacting scheme, electromagnetic field coupling is done to transfer power between the microstrip line and the radiating patch. The four most popular feed techniques used are the microstrip line, coaxial probe (both contacting schemes), aperture coupling and proximity coupling (both non-contacting schemes). 1.1.5 (a) MICROSTRIP LINE FEED In this type of feed technique, a conducting strip is connected directly to the edge of the Microstrip patch as shown in Figure 1.3. The conducting strip is smaller in width as compared to the patch and this kind of feed arrangement has the advantage that the feed can be etched on the same substrate to provide a planar structure.

Figure: 1.3 Microstrip Line Feed [7]

pg. 10


The purpose of the inset cut in the patch is to match the impedance of the feed line to the patch without the need for any additional matching element. This is achieved by properly controlling the inset position. Hence this is an easy feeding scheme, since it provides ease of fabrication and simplicity in modeling as well as impedance matching. However as the thickness of the dielectric substrate being used, increases, surface waves and spurious feed radiation also increases, which hampers the bandwidth of the antenna. The feed radiation also leads to undesired cross polarized radiation. 1.1.5 (b) COAXIAL FEED The Coaxial feed or probe feed is a very common technique used for feeding Microstrip patch antennas. As seen from Figure 1.4, the inner conductor of the coaxial connector extends through the dielectric and is soldered to the radiating patch, while the outer conductor is connected to the ground plane.

Figure: 1.4 Probe fed Rectangular Microstrip Patch Antenna [7]

The main advantage of this type of feeding scheme is that the feed can be placed at any desired location inside the patch in order to match with its input impedance. This feed method is easy to fabricate and has low spurious radiation. However, a major disadvantage is that it provides arrow bandwidth and is difficult to model since a hole has to be drilled in the substrate and the connector protrudes outside the ground plane, thus not making it completely planar for thick substrates (h >0.02λo). Also, for thicker substrates, the increased probe length makes the input impedance more inductive, leading to matching problems [11]. It is seen above that for a thick dielectric substrate,which provides broad bandwidth, the microstrip line feed and the coaxial feed suffer from numerous disadvantages. The non-contacting feed techniques which have been discussed below, solve these issues. 1.1.5 (c) APERTURE COUPLED FEED In this type of feed technique, the radiating patch and the microstrip feed line are separated by the ground plane as shown in Figure 1.5. Coupling between the patch and the feed line is made through a slot or an aperture in the ground plane.

Figure: 1.5 Aperture coupled feed [7]

pg. 11


The coupling aperture is usually centered under the patch, leading to lower cross-polarization due to symmetry of the configuration. The amount of coupling from the feed line to the patch is determined by the shape, size and location of the aperture. Since the ground plane separates the patchand the feed line, spurious radiation is minimized. Generally, a high dielectric material is used for bottom substrate and a thick, low dielectric constant material is used for the top substrate to optimize radiation from the patch . The major disadvantage of this feed technique is that it is difficult to fabricate due to multiple layers, which also increases the antenna thickness. This feeding scheme also provides narrow bandwidth. 1.1.5 (d) PROXIMITY COUPLING This type of feed technique is also called as the electromagnetic coupling Scheme. As shown in figure 1.5, two dielectric substrates are used such that the feed line is between the two substrates and the radiating patch is on top of the upper substrate. The main advantage of this feed technique is that it eliminates spurious feed radiation and provides very high bandwidth. Due to overall increase in the thickness of the microstrip patch antenna. This scheme also provides choices between two different dielectric media, one for the patch and one for the feed line..

Figure: 1.6 Proximity Feed Technique [7]

1.1.6 METHODS OF ANALYSIS The preferred models for the analysis of microstrip patch antennas are the transmission line model, cavity model, and full wave model (which include primarily integral equations /Moment Method). The transmission line model is the simplest of all and it gives good physical insight but it is less accurate. The cavity model is more accurate and gives good physical insight but is complex in nature. The full wave models are extremely accurate, versatile and can treat single elements, finite and infinite arrays, stacked elements, arbitrary shaped elements and coupling. These give less insight as compared to the two models mentioned above and are far more complex in nature. 1.1.6 (a) TRANSMISSION LINE MODEL This model represents the microstrip antenna by two slots of width W and height h, separated by a transmission line of length L. [7] The microstrip is essentially a non-homogeneous line of two dielectrics, typically the substrate and air.

Figure: 1.7 Microstrip line [7]

pg. 12

Figure: 1.8 Electric field lines [7]


Hence, as seen from Figure 1.7, most of the electric field lines reside in the substrate and parts of some lines in air. As a result, this transmission line cannot support pure transverse-electromagnetic (TEM) mode of transmission, since the phase velocities would be different in the air and the substrate. Instead, the dominant mode of propagation would be the quasiTEM mode. Hence, an effective dielectric constant (εeff) must be obtained in order to account for the fringing and the wave propagation in the line. The value of εeff is slightly less then εr because the fringing fields around the periphery of the patch are not confined in the dielectric substrate but are also spread in the air as shown in Figure 1.7 above. The expression for εreff is given by Balanis [7] as:

------1.1.6 (a) where εreff = effective dielectric constant εr h

= dielectric constant of substrate

= height of dielectric substrate

W = width of the patch Consider Figure 1.8 above, which shows a rectangular microstrip patch antenna of length L, width W resting on a substrate of height h. The co-ordinate axis is selected such that the length is along the x direction, width is along the y direction and the height is along the z direction.

Figure: 1.9 Microstrip patch antenna [7]

In order to operate in the fundamental TM10 mode, the length of the patch must be slightly less than λ/2 where λ is the wavelength in the dielectric medium and is equal to λo/√εreff where λo is the free space wavelength. The TM10 mode implies that the field varies one λ/2 cycle along the length, and there is no variation along the width of the patch. In the Figure 1.10 (a) shown below , the microstrip patch antenna is represented by two slots, separated by a transmission line of length L and open circuited at both the ends. Along the width of the patch, the voltage is maximum and current is minimum due to the open ends. The fields at the edges can be resolved into normal and tangential components with respect to the ground plane.[7]

pg. 13


Figure: 1.10(a) Top view

Figure: 1.10(b) bottom view

It is seen from Figure 1.10 (b) that the normal components of the electric field at the two edges along the width are in opposite directions and thus out of phase since the patch is λ/2 long and hence they cancel each other in the broadside direction. The tangential components (seen in Figure 1.10 (b)) , which are in phase, means that the resulting fields combine to give maximum radiated field normal to the surface of the structure. Hence the edges along the width can be represented as two radiating slots, which are λ/2 apart and excited in phase and radiating in the half space above the ground plane .The fringing fields along the width can be modeled as radiating slots and electrically the patch of the microstrip antenna looks greater than its physical dimensions. The dimensions of the patch along its length have now been extended on each end by a distance ΔL, which is given empirically by

---------1.1.6 (b)

where

--------1.1.6 (a) The effective length of the patch Leff now becomes Leff = L +2ΔL

--------1.1.6 (c)

For a given resonance frequency fr , the effective length Leff is given by [4] as

--------1.1.6 (d) For a rectangular microstrip patch antenna , the resonance frequency for any TM mn is given as

pg. 14


fo =

[( )2 + ( )2]1/2

-1.1.6 (e)

where m and n are modes along L and W respectively. for efficient radiation , the width W is given as

--------1.1.6 (f)

Where C is the free space velocity of light 1.2 METAMATERIAL The advent of micro system technologies and nanotechnologies enabled breakthroughs in many different areas of science and technology, offering functionalities well beyond the natural ones. It enabled structuring of materials for electromagnetic and optical applications in manners previously unimaginable. Among probably the best known examples of novel electromagnetic structures are photonic crystals and the negative refractive index metamaterials, popularly known as „left-handed‟ materials [17]. These enabled extension of the operation of passive and active elements for microwave and optical applications beyond the limits previously deemed impossible. Another result was an extreme miniaturization of components, sometimes even three to four orders of magnitude. Metamaterials are artificial materials engineered to provide properties which "may not be readily available in nature".These materials usually gain their properties from structure rather than composition, using the inclusion of small inhomogeneities to enact effective macroscopic behavior. Negative refractive index metamaterials (NRM) are artificial composites, characterized by subwavelength features and negative effective value of refractive index. These materials were theorically predicted in 1968 by Vaselago [4]. With the arrival of micro- and nanofabrication, new possibilities opened for practical implementation of different metamaterials and the field became intensely studied by a number of research teams. Extremely influential were seminal texts by Pendry . A further boost to the field came when the existence of NRM was experimentally confirmed by Smith, Shelby [11]. The applicability of NRM for lensing which avoids the diffraction limit by utilizing both periodic and evanescent electromagnetic waves, as proposed by Pendry in 2000 [10] even further increased the interest for NRM. The field continued to expand owing to the fact that the Maxwell equations are scalable, thus practically the same strategies can be used for the microwave and the optical range, including the transmission line approach. Today the number of the teams studying NRM and the number of published treatises on this topic are both increasing exponentially. The aim of this paper is to present a comprehensive review of the state of the art in the rapidly expanding field of NRM. We examine electromagnetic and physics of these materials, focusing our attention to some issues which may appear counter-intuitive. We systematize the most important approaches to the design and application of the NRM. 1.2.1 NEGATIVE EFFECTIVE REFRACTIVE INDEX The complex refractive index of a given medium is defined as the ratio between the speed of an electromagnetic wave through that medium and that in vacuum and can thus be written as n = με, where μ is complex relative magnetic

pg. 15


permeability and ε complex relative dielectric permittivity. If both ε and μ are negative in a given wavelength range, this means that we may write μ = |μ| exp( iπ) and in an equivalent fashion ε = |ε| exp( iπ). It follows that

= = -

-------- 1.2.1 (a)

i.e. the refractive index of a medium with simultaneously negative μ and ε must be negative. Since no known material inherently possesses negative permeability and permittivity, NRM metamaterial is a composite of two materials which individually show ε<0 and μ<0. This raises a question when it is permissible to describe such a composite as a medium with negative effective index. 1.2.2 DESIGN STRATEGIES FOR NRM It is known that highly conductive metals with permittivity dominated by plasma-like behavior (as described by Drude model) show negative dielectric permittivity in a narrow range at UV/visible frequencies. However, typical materials with negative permeability or negative permittivity are composites consisting of a large number of the basic building blocks described as unit cells (to retain correspondence with single crystals of natural materials). These building blocks are also referred to as electric (ε<0) or magnetic (μ<0) „particles‟ and are often composed of dielectric with metal inclusions. The characteristic dimensions of NRM particles have to be much smaller than the operating wavelength (in order for the effective medium approach to be applicable), but still macroscopic, i.e. much larger than the atomic or ionic dimension of their constituent materials. All NRM structures fabricated until now were highly dispersive and dissipative. The main cause of dissipation is large absorption due to conduction losses in the metal parts of the NRM. The main structures utilized to obtain NRM include thin metallic wires, metal cylinders, „Swiss roll‟ structures, split ring and complementary split ring resonators (SRR), omega structures, broadside-coupled or capacitivelly loaded SRRs, capacitivelly loaded strips, space-filling elements, etc. Of these, only the most important ones will be presented here. 1.2.2 (a) THIN METALLIC WIRES Thin metallic wires were described as one of the earliest structures with negative permittivity. The media with embedded thin metallic wires as an artificial dielectrics for microwave applications were reported in 1953 [3].The structure with ε < 0 described by Pendry consists of a square matrix of infinitely long parallel thin metal wires embedded in dielectric medium

Figure: 1.11 Metallic wire mesh with negative dielectric permittivity [3]

pg. 16


In the situation shown in Fig. 1.11 the medium is vacuum or air, the unit cell length is a and the radius of a single wire is r<
----------------

1.2.2 (i)

The effective dielectric permittivity can be written as

----------------1.2.2 (ii) i.e. it becomes negative for ω<ωp. The approximate value at the right-hand side of the above expression is valid if conductance ζ→∞. 1.2.2 (b) ‘SWISS ROLL’ STRUCTURES The induced currents in a particle (both real and displacement ones) contribute to its effective magnetization through their magnetic moments. This contribution is non-negligible if at the same time their electric polarizability is small. For instance, if the effective permeability of the structure of metal cylinders is considered, similar to that shown in Fig. 1.12, one obtains that its effective permeability cannot reach negative values. However, the introduction of capacitive elements into the structure furnishes μ<0. This can be practically done by rolling up a metal sheet into spiral coils which assume the form of a cylinder (Fig. 1.12). This is the popularly know Swiss roll structure.[8]

Figure: 1.12 swiss roll structure[8]

The sheets in the Swiss roll coils are separated by an insulator with a thickness d. If the number of coils is N and their per unit length resistance is ρ, the effective μ becomes

----- 1.2.2 (iii) Swiss roll structures are especially convenient for low frequency operation.

pg. 17


1.2.2 (c) SPLIT RING RESONATORS For decades, and starting in the early 1950s, different ring or ring-like structures with negative permeability were of interest as building blocks for artificial chiral materials in microwave. A split ring was described in this context in the textbook by Schelkunoff and Friis [1]. A double split ring resonator (SRR) (Fig1.13) is a highly conductive structure in which the capacitance between the two rings balances its inductance. A time-varying magnetic field applied perpendicular to the rings surface induces currents which, in dependence on the resonant properties of the structure, produce a magnetic field that may either oppose or enhance the incident field, thus resulting in positive or negative effective μ. In other words, the operation of a SRR represents an 'over-screened, under-damped' response of material to electromagnetic stimulation. For a circular double split ring resonator (Fig. 1.13) in vacuum and with a negligible thickness the following approximate expression is valid .

----------

1.2.2 (iv)

Where a is unit cell length and ζ is electrical conductance.

Figure: 1.13 (a) circular structure [1]

(b) square structure [1]

Dark: thin metal film

Figure: 1.14 frequency dependence of effective permittivity for a split ring resonator. Shaded area denotes negative μ region.[1]

pg. 18


The shape of the frequency dependence of є is shown in Fig. 1.14 It can be seen that there is a narrow frequency range where the effective permeability is below zero. The resonant frequency (for which μeff→±∞) Split ring resonator is probably the most often used and analyzed negative permeability building block for the NRM. 1.2.2 (d) COMPLEMENTARY SRR Structures complementary to double split rings were designed and produced by applying the Babinet principle to the split rings [14]. In this way structures with apertures in metal surface are obtained, as shown in Fig. 1.15 These complementary split rings (CSRR) [14] create negative ε instead of μ in a narrow range near the resonance frequency.

Figure: 1.15 (a) circular structure

(b) square structure

Dark : thin metal film

1.3 ORGANIZATION OF THE THESIS This topic “Design & Analysis of Microstrip patch antenna Using Metamaterial” covers all the work which I have done in this thesis.In the chapter two of this thesis The basic design of microstrip patch antenna is covered. Chapter three of this thesis covers design and analysis of microstrip patch antenna loaded with complementary split ring resonators which achieves size reduction as well as keeping the bandwidth intect. In the chapter four of this thesis the design of compact dual frequency wide band circular patch antenna with U slot is carried out and analysed. By comparision of return loss of simple circular patch antenna and circular patch antenna with U slot, it is evident that the bandwidth is greatly enhanced and there is an scope of future works in this area. In chapter five the concluding remarks and suggestion for further work of all the thesis is presented.

pg. 19


CHAPTER TWO DESIGN AND ANALYSIS OF MICROSTRIP PATCH ANTENNA 2.1 DESIGN PROCEDURE: Based on the simplified formulation that has been described, a design procedure is outlined which leads to practical design of rectangular microstrip patch antennas. The procedure assumes that the specified information includes the dielectric constant of the substrate (εr), the resonant frequency (fr) , the height of the substrate (h) and the loss tangent (δ).

SPECIFY :-

The software used to model and simulate the Microstrip patch antenna is Zeland Inc‟s IE3D. IE3D is a fullwave electromagnetic simulator based on the method of moments. It analyzes 3D and multilayer structures of general shapes. It has been widely used in the design of MICs, RFICs, patch antennas, wire antennas, and other RF/wireless antennas. It can be used to calculate and plot the S11 parameters, VSWR, current distributions as well as the radiation patterns.

Figure: 2.1 Simulated geometry of patch antenna

Length of the patch antenna = 54.775 mm width of the patch antenna

= 70.00 mm

Resonant frequency

= 1.3 GHz

pg. 20


2.1.1 SUMMARY OF DESIGN PARAMETERS

Frequency

1.3GHz

Length

54.775 mm

Width

70 mm

Cut width

5.009 mm

Cut depth

9.9 mm

Path length

44.9 mm

Path width

3.009 mm

Return loss

-24.16 dB

VSWR

1.132

Figure: 2.2 Photograph of patch antenna (top layer)

. Figure: 2.3 Photograph of patch antenna (bottom layer)

pg. 21


Figure: 2.4 current distributions

The 3D current distribution plot gives the relationship between the co-polarization (desired) and cross-polarization (undesired) components. Moreover it gives a clear picture as to the nature of polarization of the fields propagating through the patch antenna. figure 2.4 clearly shows that the patch antenna is linearly larized.

Figure: 2.5 Return loss Vs freq of patch antenna

Figure: 2.6 VSWR Vs freq of patch antenna

pg. 22


Figure: 2.7 Comparative Graph of Return loss with Frequency

Return loss is measured using a spectrum analyzer. Numerical calculations of the Return loss is performed using the method of moments (mom) based electromagnetic solver IE3D commercial software. Measured and simulated return losses are presented in Figure 2.6(a) & 2.6(b), which exhibits a good agreement. Some minor discrepancies between the measured and simulated return loss values can be attributed to some impedance mismatches as a result coax-to-microstrip transitions connector side and also imperfections in fabrication process.

2.1.2 CONCLUDING REMARKS A simple microstrip patch antenna has been designed and fabricated as shown in Figure 2.2 & 2.3. it is clear that from figure 2.5 and fig 2.7 the return loss at the frequency of operation 1.3 GHz is more than -20 dB . subsection 2.1.1 shows the summary of design parameters. By varying the cutwidth and cutdepth and of course by using the optimization method in IE3D simulation process, very good return loss up to -45 dB can be achieved. However this design suffers from the difficulties of narrow bandwidth and larger size which could be alleviated by using defected ground structure and artificially structured metamaterial in the subsequent chapter.

pg. 23


CHAPTER THREE MICROSTRIP PATCH ANTENNA LOADED WITH COMPLEMENTARY SRR 3.1 INTRODUCTION We present characteristics of microstrip patch antennas on metamaterial substrates loaded with complementary split-ring resonators (CSRRs). The proposed antenna utilizes CSRRs in the ground plane altering the effective medium parameters of the substrate. To characterize the performance of the CSRR loaded microstrip antenna, the metamaterial substrate has been modeled as an effective medium with extracted constitutive parameters. Simulation results were verified by experimental results. The experimental results confirm that the CSRR loaded patch antenna achieves size reduction and maintaining the same bandwidth as well. Metamaterials [13] are finding numerous applications for novel antennas. One such application is the use of artificial materials for compact antennas. Miniaturization of microstrip antennas has been attempted for a long time using various different methods. Most popular and traditional way would be to use a high permittivity substrate decreasing the guided wavelength in the substrate, so that the overall antenna size is reduced [4]. However, this approach has a drawback resulting in the tendency for more of the energy delivered to the antenna to be trapped in substrates with high permittivities, which eventually decreases the antenna impedance bandwidth. To overcome the drawbacks of the patch antenna on a high permittivity substrate, several remedies have been proposed using artificial structures in conjunction with the patch element [6], [19]. In this chapter, we propose a new design approach to the realization of compact antennas with improved impedance bandwidth using an artificial substrate based on complementary split ring resonators (CSRRs) and present the simulated and measured characteristics of the designed antenna. The characteristics of split-ring resonators (SRRs) have been already studied by several groups [12], [18]. In complementary structures, due to the fact that the electric boundary conditions on the metal are exchanged with magnetic ones, the structure becomes effectively dual. thus, the CSRRs electromagnetic behavior is essentially an electric dipole excited by an axial electric field exhibiting similar propagation properties as an effective negative є medium [14],[15]. We investigate a microstrip patch antenna on a metamaterial substrate with CSRRs employed in the ground plane, and examine the resonant frequency, impedance bandwidth, and radiation characteristics using the effective medium approach. The comparison of the impedance bandwidth between the microstrip patch antenna on a conventional high permittivity substrate and with the CSRR loaded metamaterial substrate is presented. The experimental results demonstrated that a size reduction is possible for a microstrip antenna without sacrificing the bandwidth by using the metamaterial substrate based on CSRRs. 3.1.1 PROPOSED ANTENNA CONFIGURATION Figure 3.1 shows the geometry of the CSRR and antenna configuration. In the proposed antenna, the solid metal ground plane is replaced with a ground plane with periodically etched CSRRs. [20]

Figure:3.1 Configuration of the metamaterial substrate antenna

pg. 24


.

Figure:3.2 Comparison of simulated return losses between the conventional antenna and the CSRR loaded antenna for different substrate permitivities (eps).[20]

Figure 3.2 shows the simulated return losses for three different antennas (patch antennas on conventional dielectric substrates (єr=3 and 6) and on the metamaterial substrate loaded with CSRRs (є r=3 ). It has been noted that the CSRR loaded antenna with єr=3 achieves the same reduction in the resonant frequency as the conventional antenna with substrate material having a twice higher permittivity. Moreover, the bandwidth at the operating frequency is wider than the original operating frequency. The increased bandwidth can be attributed to the reduced Q of the substrate due to the energy leakage through the aperture in the CSRR. 3.1.2 DESIGN ANALYSIS OF A CSRR LOADED PATCH ANTENNA

Figure: 3.3 simulated Geometry of patch loaded with CSRR.

Unit cell parameters: R1 = 8 mm R2 = 5 mm C = 2 mm D = 1 mm

pg. 25

Figure: 3.4 Unit cell CSRR


Figure: 3.5 (a) simulated Geometry of patch loaded with CSRR(upper layer)

Upper layer parameters: Length

= 54.775 mm

Width

= 70 mm

Cut width = 5.009 mm Cut depth = 9.9 mm

Figure:3.5(b) simulated Geometry of patch loaded with CSRR(bottom layer)

CSRR array parameters: Length of the CSRR array = 86 mm width of the CSRR array = 50 mm

pg. 26


Figure: 3.6(a) Photograph of the fabricated antenna (Upper layer) (Upper layer)

Figure: 3.6(b) Photograph of the fabricated antenna

Figure: 3.7(a) Photograph of the fabricated antenna (bottom layer)

Figure 3.7(b) Photograph of the fabricated antenna (bottom layer)

Return loss is measured using a spectrum analyzer. Numerical calculations of the Return loss is performed using the Method of Moments (Mom) based electromagnetic solver IE3D commercial software. Measured and simulated return losses are presented in Figure 2.6(a) & 2.6(b), which exhibits a good agreement. Some minor discrepancies between the measured and simulated return loss values can be attributed to some impedance mismatches as a result coax-to-microstrip transitions connector side and also imperfections in fabrication process.

pg. 27


Figure: 3.8 Return loss Vs freq of a normal patch antenna

Figure: 3.9 Return loss vs frequency of CSRR loaded patch antenna

pg. 28


Figure: 3.10 Angle Vs freq showing phase change at 1.4 GHz of a metamaterial(CSRR) loaded patch antenna.

Figure: 3.11 comparative graph of return loss Vs frequency

3.13 CONCLUDING REMARKS A compact microstrip antenna with an improved bandwidth using a metamaterial substrate based on complimentary split ring resonators has been presented. For the characterization of the microstrip antennas on metamaterial substrates, the effective medium approach was employed. The new design help achieve the reduction of the antenna size and the improvement of the bandwidth for microstrip patch antennas. The results presented in this chapter are promising for the design of compact antennas achieving a size reduction without having to sacrifice the antenna bandwidth, which makes the antenna useful for various applications. pg. 29


CHAPTER FOUR COMPACT DUAL FREQUENCY WIDE BAND CIRCULAR PATCH ANTENNA WITH U-SLOT 4.1 INTRODUCTION In this chapter, we have reported the performance of a compact inset-feed circular patch antenna with U-shaped slot and its performance is compared with that of a simple circular patch antenna designed under identical conditions. The radiation properties of a circular patch antenna with U slot designed on glass epoxy FR-4 substrate are obtained and compared with that of a normal circular patch antenna designed under identical conditions. The modified antenna not only resonates at two different frequencies (2.567-2.733 GHz) but also presents marked improvement in the bandwidth (3 dB bandwidth 400 MHz). return loss and VSWR of antenna as a function of frequency are simulated using conventional software Zeeland‟s IE3D method of moment and has been compared with the network analyzer measured value of return loss Vs frequency. while radiation patterns of the modified structure are simulated to present in this paper.

4.1.1 PROPOSED ANTENNA GEOMETRY Geometry and inset feed arrangement of a circular patch antenna with U-shaped slot is shown in figure4.I. The circular patch antenna with radius 'r' = 1.4 cm is designed on a glass epoxy FR-4 substrate having substrate thickness 'h' = 1.6mm, relative dielectric constant = 4.4 and loss tangent tan β= 0.02. The patch lie in the XY plane over a large rectangular ground plane. the resonance frequency of a simple circular patch antenna for TM10 mode of excitation is 2.877 GHz while measured results with this antenna indicate that antenna is resonating at single frequency 2.86GHz. These two frequency values are quite close to each other. The measured VSWR of this antenna is 1.15. These results indicate good matching of antenna geometry with the feed network.

Figure: 4.1 Proposed Geometry of circular patch antenna with U slot [5]

pg. 30


4.1.2 DESIGN OF CIRCULAR PATCH ANTENNA The circular patch antenna with radius 'r' = 1.4 cm is designed on a glass epoxy FR-4 substrate having substrate thickness 'h' = 1.6 mm, relative dielectric constant = 4.4 and loss tangent tan β = 0.02. The patch lie in the XY plane over a large rectangular ground plane

< ------2.8 c.m ------------ > Feed point : 3,-5.5 Figure:4.2 simple circular patch antenna

Figure: 4.3 Variation of return loss with frequency for a circular patch antenna

pg. 31


4.1.3 DESIGN OF CIRCULAR PATCH ANTENNA WITH U SLOT : A U slot has been made on the same circular patch antenna with radius 'r' = 1.4 cm is designed on a glass epoxy FR-4 substrate having substrate thickness 'h' = 1.6mm, relative dielectric constant = 4.4 and loss tangent tan β = 0.02. The patch lie in the XY plane over a large rectangular ground plane. The structure is simulated by IE3D using method of moment technique having feed point at 3,-5.5 and its response of return loss Vs frequency is analyses.

-----> I 5 mm I <---

Figure: 4.4 circular patch antenna with U slot made at the patch.

Upper structure Figure: 4.5 photograph of U slot circular patch antenna (top layer)

Figure: 4.6 photograph of U slot circular patch antenna (bottom layer)

pg. 32


Figure: 4.7 Variation of return loss with frequency for a circular patch antenna with U slot made at the patch.

Figure 4.8 comparative graph of return loss Vs frequency

Figure 4.9 variation of VSWR with frequency

pg. 33


On application of U-shaped slot in the circular patch geometry with dimensions LI = 1.05cm, L2 = 0.95cm, WI = 0.5cm, and W2 = 0.l cm, the antenna now resonates at two different frequencies (2.568 and 2.677 GHz) which are smaller than that obtained with a simple circular patch antenna. The variations in return loss and VSWR with frequency are shown in figures 4.7 , 4.8 and 4.9. VSWR of antenna is close to unity at the frequency of interest as shown in the figure 4.9 which again confirms excellent matching of this antenna with feed network. 4.1.4 CONCLUDING REMARKS The return loss of a circular patch antenna with U shaped slot is investigated in free space and is compared with that of a simple circular patch antenna excited under similar conditions. The modified antenna resonates at two different frequencies with improved bandwidth. The two resonance frequencies are in the lower band of frequencies allotted for Wi-Fidelity systems by IEEE 802.16 working group. The verification of other results through experimental results is still going on.

pg. 34


CHAPTER FIVE CONCLUSION AND SUGGESTIONS FOR FUTURE WORK 5.1 CONCLUDING REMARKS Microstrip patch antenna can provide printed radiating structure, which are electrically thin, lightweight and low cost, is a relatively not too old. The development of system such as Satellite communication, highly sensitive radar, radio altimeters and Missiles systems needs very light weight antenna which can be easily attached with the systems and not make the system bulky. These requirements are main factors to the development of the microstrip patch antenna. By doing this we can get required results. Rectangular and circular microstrip patch antenna are most common and very easy to analysis but to enhance their bandwidth, and to achieve multiband operation we need to make some slots on the patch and to work on defected ground structure, defected microstrip structure and meta-material. 5.2 SUGGESTION FOR FUTURE WORKS In future also, as per requirement many new shapes can replace the conventional shapes .There are many shapes in the field of microstrip patch antenna .A design of slots on the patch and making defected structure in the ground plane for improving the bandwidth as well as achieving the multiband operation which is the part of this project is very good for future aspects. All works has been performed in the thesis with the IE3D simulation software. An interesting feature of a CSRR loaded patch antenna which would be my future area of work is the zeroth order resonance [16], which could be observed by simulating the CSRR loaded patch antenna at different resonant frequency. Observing zeroth order resonance could be utilized for designing multiband antennas. As we have seen the utility of U slot antenna for achieving the wideband as well as multiband operation. This advantage can further be enhanced by using double U slot antenna.

pg. 35


REFERENCES [1] S. A. Schelkunoff, H.T.Friss, Antennas: Theory and Practice, New York: John Willy & Sons, 1952. [2 ] G. A. Deschamps, “ Microstrip Microwave Antennas”, presented at Third USAF symposium on Antennas, 1953. [3] J. Brown, “Artificial dielectrics,” Progress in Dielectrics, vol. 2, pp. 195–225, 1960. [4] V. G. Veselago,“The electrodynamics of substances with simultaneously negative values of epsilon and mu,” Sov. Phys. Uspekhi, 10, pp. 509-514, 1968. [5] Y.T. Lo, Theory and experiment on microstrip antennas, IEEE Trans.Antennas Propag 27, pp.137–145. 1979 [6] J.S. Colburn and Y. Rahmat-Samii, patch antennas on externally perforated high dielectric permittivity material, electron Lett. 31,pp.1710–1712. 1995 [7] C. A. Balanis, “Antenna Theory, Analysis and Design,” John Wiley & Sons, New York, 1997. [8] J. B. Pendry, A. J. Holden, D. J. Robbins, and W. J. Stewart, “Magnetism from conductors and enhanced nonlinear phenomena,” IEEE Trans. Microw. Theory Tech. 47, pp. 2075-2081, 1999. [9] R Garg, P Bhartia, I Bahl, and A. Lttipiboon, Microstrip antenna design handbook, Artech House, 2000. [10] J. B. Pendry, “Negative Refraction Makes a Perfect Lens,” Phys. Rev. Lett. 85,pp.3966-3969, 2000. [11] A. Shelby, D. R. Smith, S. Schultz, “Experimental verification of a negative index of refraction,” Science, 292, pp. 77–79, 2001. [12] R. Marques, F. Mesa, J. Martel, and F. Median, Comparative analysis of edge and broadside coupled split ring resonators for metamaterial design-Theory and experiment, IEEE Trans. Antennas Propag.51,pp 2572–2581.2003, [13] D.R. Smith, J.B. Pendry, and M.C.K.Wiltshire, Metamaterials and negative refractive index, Science, 305, pp.788–792. 2004, [14] F. Falcone, T. Lopetegi, M. A. G. Laso, J. D. Baena, J. Bonache, M. Beruete, R. Marques, F. Martin, M. Sorolla, “Babinet Principle Applied to the Design of Metasurfaces and Metamaterials,” Phys. Rev. Lett. 93, 197401, 2004. [15] F. Falcone, T. Lopetegi, J.D. Baena, R. Marques, F. Martin, and M. Orolla, Effective Negative є stopband microstrip lines based on complimentary split ring resonators, IEEE Microwave Wireless Compon. Lett. 14,pp.280–282. 2004 [16 ] A. Sanada, M. Kimura, I. Awai, C. Caloz, and T. Itoh, A Planar Zeroth-Order Resonator Antenna Using A Left-handed TransmissionLine, 34th European Microwave Conference, Amsterdam, 2004. [17] S. A. Ramakrishna, “Physics of negative refractive index materials,” Rep. Prog. Phys. 68, pp. 449–521, 2005. [18] J.D. Baena, J. Bonache, F. Martin, R.M. Sillero, F. Falcone, T. Lopetegi, M.A.G. Laso, J. Garcia-Garcia, I. Gil, M.F. Portillo, and M.Sorolla, Equivalent-circuit models for split-ring resonators and complementary split-ring resonators coupled to planar transmission lines, IEEE Trans. Antennas Propag.53, 1451–1461. 2005 [19] P.M.T. Ikonen, S.I. Maslovski, C.R. Simovski, and S.A. Tretyakov, on artificial magnetodielectric loading for improving the impedance bandwidth properties of microstrip antennas, IEEE Trans. Antennas Propag. 54, pp.1654–1662. 2006 [20] Yoonjae Lee and Yang Hao, “Characterization of microstrip patch antennas on metamaterial Substrates loaded with complementary split-ring Resonators” Wiley Periodicals, Inc. Microwave Opt Technol. Lett. 50, pp.2131–2135, 2008.

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A B O U T T H E AU T H O R Sunil Kumar Thakur is presently working as Senior Scientific Officer In Director General Quality Assurance, Department of Defense production, Ministry Of Defense (India) since Jul 2012. Till Jul 2012 he had 19 years of working experience in Indian Air Force wherein he has been looking after induction of Surface To Air Missile Systems in Indian Air Force and Production, Planning, Execution and its Monitoring of Missiles and Radars systems of Russian and Indio-western origin. He has received his Bachelor of Engineering Degree in Electronics & Communication Engineering from Institution of Engineers (India) and ME Degree in Communication Control and Networking from the University of RGPV Bhopal, MP. He has presented 03 papers in National Conference and one paper in International Conference in the topic of Microstrip Patch Antenna.

pg. 37


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