Este Documento Ha sido descargado desde la Web más completa en todo tipo de ebooks y Tutoriales.
Si deseas más información o libros, entonces entonces ingresa a: http://www.enigmaelectronica.tk http://www.foroenigma.tk Y podrás descargar muchas aplicaciones útiles.
Libros Manuales Tutoriales Cursos Programas Música Películas Grupo Enigma Electrónica Enigma Team
Si algún Archivo Requiriera de Contraseña de acceso siempre será: www.enigmaelectronica.tk
Este Documento Ha sido descargado desde la Web más completa en todo tipo de ebooks y Tutoriales.
Si deseas más información o libros, entonces entonces ingresa a: http://www.enigmaelectronica.tk http://www.foroenigma.tk Y podrás descargar muchas aplicaciones útiles.
Libros Manuales Tutoriales Cursos Programas Música Películas Grupo Enigma Electrónica Enigma Team
Si algún Archivo Requiriera de Contraseña de acceso siempre será: www.enigmaelectronica.tk
SWITCHMODE
t
Power Supplies
Reference Manual and Design Guide
SMPSRM/D Rev. Rev. 3B, July-2002 July-200 2
© SCILLC, 2007 Previous Edition © 2002 “All Rights Reserved''
SMPSRM
ON Semiconductor and are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particul ar purpose, nor does SCILLC assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages. “Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operatin g parameters, including “Typicals” must be validated for each customer application by customer 's technical experts. SCILLC does not convey any license under its patent rights nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur. Should Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, c osts, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part. SCILLC is an Equal Opportunity/Affirmative Action Employer. This literature is subject to all applicable copyright laws and is not for resale in any manner.
PUBLICATION ORDERING INFORMATION LITERATURE FULFILLMENT: Literature Distribution Center for ON Semiconductor .O. Box 5163, Denver, Colorado 80217 USA P Phone: 303-675-2175 or 800-344-3860 Toll Free USA/Canada Fax : 303-675-2176 or 800-344-3867 Toll Free USA/Canada Email:
[email protected]
N. American Technical Support: 800-282-9855 Toll Free USA/Canada Europe, Middle East and Africa Technical Support: Phone: 421 33 790 2910 Japan Customer Focus Center Phone: 81-3-5773-3850
www.onsemi.com 2
ON Semiconductor Website: www.onsemi.com Order Literature: http://www.onsemi.com/orderlit For additional information, please contact your local Sales Representative
SMPSRM
Forward Every new electronic product, except those that are battery powered, requires converting off-line 115 Vac or 230 Vac power to some dc voltage for powering the electronics. The availability of design and application information and highly integrated semiconductor control ICs for switching power supplies allows the designer to complete this portion of the system design quickly and easily. Whether you are an experienced power supply designer, designing your first switching power supply or responsible for a make or buy decision for power supplies, the variety of information in the SWITCHMODEā™ Power Supplies Reference Manual and Design Guide should prove useful. ON Semiconductor has been a key supplier of semiconductor products for switching power supplies since we introduced bipolar power transistors and rectifiers designed specifically for switching power supplies in the mid-70's. We identified these as SWITCHMODE ™ products. A switching power supply designed using ON Semiconductor components can rightfully be called a SWITCHMODE power supply or SMPS. This brochure contains useful background information on switching power supplies for those who want to have more meaningful discussions and are not necessarily experts on power supplies. It also provides real SMPS examples, and identifies several application notes and additional design resources available from ON Semiconductor, as well as helpful books available from various publishers and useful web sites for those who are experts and want to increase their expertise. An extensive list and brief description of analog ICs, power transistors, rectifiers and other discrete components available from ON Semiconductor for designing a SMPS are also provided. This includes our newest GreenLine ™, Easy Switcher and very high voltage ICs (VHVICs), as well as high efficiency HDTMOS ® and HVTMOS® power FETs, and a wide choice of discrete products in surface mount packages. For the latest updates and additional information on analog and discrete products for power supply and power management applications, please visit our website: ( www.onsemi.com ).
MEGAHERTZ, POWERTAP, SENSEFET, SWITCHMODE, and TMOS are trademarks of Semiconductor Components Industries, LLC. HDTMOS and HVTMOS are registered trademarks of Semiconductor Components Industries, LLC. GreenLine, SMARTMOS and Motorola are trademarks of Motorola Inc.
www.onsemi.com 3
SMPSRM
Table of Contents Page
Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Linear versus Switching Power Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Switching Power Supply Fundamentals . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . The Forward-Mode Converter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . The Flyback-Mode Converter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Common Switching Power Supply Topologies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Interleaved Multiphase Converters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Selecting the Method of Control . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . The Choice of Semiconductors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Power Switches . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . The Bipolar Power Transistor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . The Power MOSFET . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Driving MOSFETs in Switching Power Supply Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . The Insulated Gate Bipolar Transistor (IGBT) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Rectifiers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . The Magnetic Components . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Laying Out the Printed Circuit Board . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Losses and Stresses in Switching Power Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Techniques to Improve Efficiency in Switching Power Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . The Synchronous Rectifier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Snubbers and Clamps . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . The Lossless Snubber . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . The Active Clamp . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Quasi-Resonant Topologies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Power Factor Correction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
5 5 5 5 7 8 13 14 16 16 16 17 18 19 19 21 21 24 25 25 27 28 29 30 32
SMPS Examples . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Integrated Circuits for Switching Power Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Suggested Components for Specific Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
35 36 37
Literature Available from ON Semiconductor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Application Notes, Brochures, Device Data Books and Device Models . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . References for Switching Power Supply Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Books . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Websites . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
56 56 58 58 59
www.onsemi.com 4
SMPSRM
Introduction
A low drop-out (LDO) regulator uses an improved output stage that can reduce Vdrop to considerably less than 1.0 V. This increases the efficiency and allows the linear regulator to be used in higher power applications. Designing with a linear regulator is simple and cheap, requiring few external components. A linear design is considerably quieter than a switcher since there is no high-frequency switching noise. Switching power supplies operate by rapidly switching the pass units between two efficient operating states: cutoff , where there is a high voltage across the pass unit but no current flow; and saturation, where there is a high current through the pass unit but at a very small voltage drop. Essentially, the semiconductor power switch creates an AC voltage from the input DC voltage. This AC voltage can then be stepped-up or down by transformers and then finally filtered back to DC at its output. Switching power supplies are much more efficient, ranging from 65 to 95 percent. The downside of a switching design is that it is considerably more complex. In addition, the output voltage contains switching noise, which must be removed for many applications. Although there are clear differences between linear and switching regulators, many applications require both types to be used. For example, a switching regulator may provide the initial regulation, then a linear regulator may provide post-regulation for a noise-sensitive part of the design, such as a sensor interface circuit.
The never-ending drive towards smaller and lighter products poses severe challenges for the power supply designer. In particular, disposing of excess heat generated by power semiconductors is becoming more and more difficult. Consequently it is important that the power supply be as small and as efficient as possible, and over the years power supply engineers have responded to these challenges by steadily reducing the size and improving the efficiency of their designs. Switching power supplies offer not only higher efficiencies but also greater flexibility to the designer. Recent advances in semiconductor, magnetic and passive technologies make the switching power supply an ever more popular choice in the power conversion arena. This guide is designed to give the prospective designer an overview of the issues involved in designing switchmode power supplies. It describes the basic operation of the more popular topologies of switching power supplies, their relevant parameters, provides circuit design tips, and information on how to select the most appropriate semiconductor and passive components. The guide also lists the ON Semiconductor components expressly built for use in switching power supplies.
Linear versus Switching Power Supplies Switching and linear regulators use fundamentally different techniques to produce a regulated output voltage from an unregulated input. Each technique has advantages and disadvantages, so the application will determine the most suitable choice. Linear power supplies can only step-down an input voltage to produce a lower output voltage. This is done by operating a bipolar transistor or MOSFET pass unit in its linear operating mode; that is, the drive to the pass unit is proportionally changed to maintain the required output voltage. Operating in this mode means that there is always a headroom voltage, Vdrop, between the input and the output. Consequently the regulator dissipates a considerable amount of power, given by (Vdrop Iload). This headroom loss causes the linear regulator to only be 35 to 65 percent efficient. For example, if a 5.0 V regulator has a 12 V input and is supplying 100 mA, it must dissipate 700 mW in the regulator in order to deliver 500 mW to the load , an efficiency of only 42 percent. The cost of the heatsink actually makes the linear regulator uneconomical above 10 watts for small applications. Below that point, however, linear regulators are cost-effective in step-down applications.
Switching Power Supply Fundamentals There are two basic types of pulse-width modulated (PWM) switching power supplies, forward-mode and boost-mode. They differ in the way the magnetic elements are operated. Each basic type has its advantages and disadvantages. The Forward-Mode Converter The forward-mode converter can be recognized by the presence of an L-C filter on its output. The L-C filter creates a DC output voltage, which is essentially the volt-time average of the L-C filter's input AC rectangular waveform. This can be expressed as: Vout [ Vin @ dutyĂcycle
(eq. 1)
The switching power supply controller varies the duty cycle of the input rectangular voltage waveform and thus controls the signal's volt-time average. The buck or step-down converter is the simplest forward-mode converter, which is shown in Figure 1.
www.onsemi.com 5
SMPSRM LO
SW
Ion
Vin
E G A T ) S L T O L V O E V D ( O I D T N E R R ) U S C P R M A O T ( C U D N I
Ioff
D
Rload
Cout
Vsat
Power Switch ON
Power Switch OFF
Power Switch ON
Power Switch OFF TIME
Vfwd
Ipk Iload Imin Power SW
Diode
Power SW
Diode TIME
Figure 1. A Basic Forward-Mode Converter and Waveforms (Buck Converter Shown)
Its operation can be better understood when it is broken into two time periods: when the power switch is turned on and turned off. When the power switch is turned on, the input voltage is directly connected to the input of the L-C filter. Assuming that the converter is in a steady-state, there is the output voltage on the filter's output. The inductor current begins a linear ramp from an initial current dictated by the remaining flux in the inductor. The inductor current is given by: iL(on) +
(Vin * Vout) t ) iinit L
0 v t v ton
clamped when the catch diode D becomes forward biased. The stored energy then continues flowing to the output through the catch diode and the inductor. The i pk and is inductor inducto r current decreases from an initial value pk given by: iL(off) + ipk *
Voutt L
0 v t v toff
(eq. 3)
The off period continues until the controller turns the power switch back on and the cycle repeats itself. The buck converter is capable of over one kilowatt of output power, but is typically used for on-board regulator applications whose output powers are less than 100 watts. Compared to the flyback-mode converter, the forward converter exhibits lower output peak-to-peak ripple voltage. The disadvantage is that it is a step-down topology only. Since it is not an isolated topology, for safety reasons the forward converter cannot be used for input voltages greater than 42.5 VDC.
(eq. 2)
During this period, energy is stored as magnetic flux within the core core of the inductor. inductor. When the power switch is turned off, the core contains enough energy to supply the load during the following off period plus some reserve energy. When the power switch turns off, the voltage on the input side of the inductor tries to fly below ground, but is
www.onsemi.com 6
SMPSRM The Flyback-Mode Converter The basic flyback-mode converter uses the same components componen ts as the basic forward-mode converter, but in a different configuration. Consequently, Consequently, it operates in a
different fashion from the forward-mode converter. The most elementary flyback-mode converter, the boost or step-up converter, is shown in Figure 2.
L D
Cout Vin
SW Ion
Ioff
Iload
Rload
Vin E G A T L ) S O T V L H O V C ( T I W S
Vflbk (Vout)
Power Switch ON
Vsat
Power Switch ON
Diode ON
Power Switch ON
Diode ON TIME
T N E R R ) U S C P R M A O T ( C U D N I
Ipk
Iload TIME
Figure 2. A Basic Boost-Mode Converter and Waveforms (Boost Converter Shown)
Again, its its operatio operation n is best underst understood ood by consi consideri dering ng the “on” and “off” periods separately. When the power switch is turned on, the inductor is connected directly across the input voltage source. The inductor inductor current then rises from zero and is given by: V t iL(on) + in L
v t v 0on
the output rectifier when its voltage exceeds the output voltage. The energy within the core of the inductor is then passed to the output capacitor. The inductor current during the off period has a negative ramp whose slope is given by:
(eq. 4)
iL(off) +
Energy is stored within the flux in the core of the inductor. i pk , occurs at the instant the power The peak current, pk switch is turned off and is given by: ipk +
VinĂton L
(Vin * Vout) L
(eq. 6)
The energy is then completely emptied into the output capacitor and the switched terminal of the inductor falls back to the level of the input voltage. Some ringing is evident during this time due to residual energy flowing through parasitic elements such as the stray inductances and capacitances in the circuit.
(eq. 5)
When the power switch turns off, the switched side of the inductor wants to fly-up in voltage, but is clamped by
www.onsemi.com 7
SMPSRM When there is some residual energy permitted to remain within the inductor core, the operation is called continuous- mode. This can be seen in Figure 3.
to a 50 percent duty cycle. There must be a time period when the inductor is permitted to empty itself of its energy. The boost converter is used for board-level (i.e., non-isolated) step-up applications and is limited to less than 100-150 watts due to high peak currents. Being a non-isolated converter, it is limited to input voltages of less than 42.5 VDC. Replacing the inductor with a transformer results in a flyback converter, which may be step-up or step-down. The transformer also provides dielectric isolation from input to output.
Energy for the entire on and off time periods must be stored within the inductor. The stored energy is defined by: EL + 0.5L @ ipk2
(eq. 7)
The boost-mode inductor must store enough energy to supply the output load for the entire switching switching period (ton + toff ). Also, boost-mode converters are typically limited E G A T ) L S O T V L H O V C ( T I W S
Vflbk (Vout) Vin Power Switch ON
Power Switch ON
Diode ON
Diode ON TIME
T N E R R ) U S C P R M A O T ( C U D N I
Vsat Ipk
TIME
Figure 3. Waveforms for a Continuous-Mode Continuous-Mode Boost Converter
Common Switching Power Supply Topologies
5. How much of the input voltage voltage is placed across the primary transformer winding or inductor? Factor 1 is a safety-related issue. Input voltages above 42.5 VDC are considered hazardous by the safety regulatory agencies throughout the world. Therefore, only transformer-isolated transformer-isolated topologies must be used above this voltage. These are the off-line applications where the power supply is plugged into an AC source such as a wall socket. Multiple outputs require a transformer-based topology. The input and output grounds may be connected together if the input voltage is below 42.5 VDC. V DC. Otherwise full dielectric isolation is required.
A topology is the arrangement of the power devices and their magnetic elements. Each topology has its own merits within certain applications. There are five major factors to consider when selecting a topology for a particular application. These are: 1. Is input-to-outpu input-to-outputt dielectric dielectric isolation isolation required required for the application? This is typically dictated by the safety regulatory bodies in effect in the region. 2. Are multiple multiple outputs required? 3. Does the prospective prospective topology topology place a reasonable reasonable voltage stress across the power semiconductors? 4. Does the prospective prospective topology topology place a reasonable reasonable current stress upon the power semiconductors?
www.onsemi.com 8
SMPSRM Factors 3, 4 and 5 have a direct affect upon the reliability of the system. Switching power supplies deliver constant power to the output load. This power is then reflected back to the input, so at low input voltages, the input current must be high to maintain the output power. Conversely, the higher the input voltage, the lower the input current. The design goal is to place as much as possible of the input voltage across the transformer or inductor so as to minimize the input current. Boost-mode topologies have peak currents that are about twice those found in forward-mode topologies. This makes them unusable at output powers greater than 100-150 watts.
Cost is a major factor that enters into the topology decision. There are large overlaps in the performance boundaries between the topologies. Sometimes the most cost-effective choice is to purposely design one topology to operate in a region that usually is performed by another. This, though, may affect the reliability of the desired topology. Figure 4 shows where the common topologies are used for a given level of DC input voltage and required output power. Figures 5 through 12 show the common topologies. There are more topologies than shown, such as the Sepic and the Cuk, but they are not commonly used.
1000 Half-Bridge ) V ( E G A 100 T L O V T 42.5 U P N I C D 10
Flyback
Non-Isolated
Full-Bridge
Full-Bridge Very High
Buck
Peak Currents
10
100
1000
OUTPUT POWER (W)
Figure 4. Where Various Topologies Are Used
www.onsemi.com 9
SMPSRM L Power Switch
+
+ D Vin
Cin
VFWD
VD 0
IPK
Vout
+ Cout
Control
IL
Feedback
0
-
TIME
Vin
TIME
ILOAD
-
IMIN
Figure 5. The Buck (Step-Down) Converter
VFLBK
VSW
L
0
+ +
SW
Control
D ON SW ON
D
Cin
Vin
D ON
VSAT
TIME Vin
Vout
Cout
IPK
IL -
ISW
0
ID
TIME
Figure 6. The Boost (Step-Up) Converter
+ VL
Control Vin
Vin TIME
0
SW D
Cin
+
L -
-ĂVout
Vout
Cout
+
Feedback
IL
ISW
0
ID
TIME
IPK
Figure 7. The Buck-Boost (Inverting) Converter VFLBK
VSAT
SW ON
VSW
TIME
0 +
Vin
D N1
Cin Control
-
Vin
N2
+ Cout
+ Vout
IPRI 0
-
SW
IPK
TIME
ISEC Feedback
0
Figure 8. The Flyback Converter
www.onsemi.com 10
TIME
SMPSRM +
LO
D
T
+ N2
N1
Cout
Cin
Vin
+
Vout -
SW
Control -
Feedback
SW ON
VSW
TIME
0 VSAT
2Vin
IPRI 0
TIME IMIN
IPK
Figure 9. The One-Transistor Forward Converter (Half Forward Converter)
SW1
D1
T
LO +
+
D2
Cout
+
SW2 Vin Cin
-
Control
-
Feedback
2Vin SW2
Vin VSW
SW1 TIME
0 VSAT IPK
IPRI
Vout
0
TIME IMIN
Figure 10. The Push-Pull Converter
www.onsemi.com 11
SMPSRM LO Ds
+ + Cout
Vout -
+ N2
XFMR
SW1
T
Cin Control
Vin
N1
SW2
C C
Feedback Vin SW1
Vin 2 SW2
VSW2 0
TIME VSAT IPK
IPRI TIME
0 IMIN
Figure 11. The Half-Bridge Converter LO Ds
+ Cout
+
-
+ XFMR Vin
Vout
N2
SW1
T
Cin
N1
Control
SW3
XFMR
C SW4
SW2 -
Vin SW Vin 2
1‐4
SW 2‐3
VSW2 0
TIME VSAT IPK
ISW2 0
TIME IMIN
Figure 12. The Full-Bridge Converter
www.onsemi.com 12
SMPSRM
Interleaved Multiphase Converters
The input and output capacitors are shared among the phases. The input capacitor sees less RMS ripple current because the peak currents are less and the combined duty cycle of the phases is greater than it would experience with a single phase converter. The output capacitor can be made smaller because the frequency of current waveform is n-times higher and its combined duty cycle is greater. The semiconductors also see less current stress. A block diagram of an interleaved multiphase buck converter is shown in Figure 13. This is a 2-phase topology that is useful in providing power to a high performance microprocessor.
One method of increasing the output power of any topology and reducing the stresses upon the semiconductors, is a technique called interleaving. Any topology can be interleaved. An interleaved multiphase converter has two or more identical converters placed in parallel which share key components. For an n-phase converter, each converter is driven at a phase difference of 360/ n degrees from the next. The output current from all the phases sum together at the output, requiring only Iout / n amperes from each phase.
+ +
VIN
CIN
-
S A1
L A
S A2
VFDBK GATEA1 Control
+
GATEA2 +
GND
GATEB2
CFA
GATEB1
COUT SB1
LB
CFB CS5308
SB2
Current Feedback A Current Feedback B Voltage Feedback
Figure 13. Example of a Two-Phase Buck Converter with Voltage and Current Feedback
www.onsemi.com 13
VOUT -
SMPSRM
Selecting the Method of Control
select the one that is wanted. Table 1 summarizes the features of each of the popular methods of control. Certain methods are better adapted to certain topologies due to reasons of stability or transient response.
There are three major methods of controlling a switching power supply. There are also variations of these control methods that provide additional protection features. One should review these methods carefully and then carefully review the controller IC data sheets to Table 1. Common Control Methods Used in ICs Control Method Voltage-Mode
Current-Mode Hysteric Voltage
OC Protection
Response Time
Preferred Topologies
Average OC
Slow
Forward-Mode
Pulse-by-Pulse OC
Slow
Forward-Mode
Intrinsic
Rapid
Boost-Mode
Hysteretic
Rapid
Boost & Forward-Mode
Average
Slow
Boost & Forward-Mode
Voltage-mode control (see Figure 14) is typically used for forward-mode topologies. In voltage-mode control, only the output voltage is monitored. A voltage error signal is calculated by forming the difference between Vout (actual) and Vout(desired). This error signal is then fed into a comparator that compares it to the ramp voltage generated by the internal oscillator section of the control IC. The comparator thus converts the voltage error signal into the PWM drive signal to the power switch. Since the only control parameter is the output voltage, and there is inherent delay through the power circuit, voltage-mode control tends to respond slowly to input variations. Overcurrent protection for a voltage-mode controlled converter can either be based on the average output current or use a pulse-by-pulse method. In average overcurrent protection, the DC output current is monitored, and if a threshold is exceeded, the pulse width of the power switch is reduced. In pulse-by-pulse overcurrent protection, the peak current of each power switch “on” cycle is monitored and the power switch is
instantly cutoff if its limits are exceeded. This offers better protection to the power switch. Current-mode control (see Figure 15) is typically used with boost-mode converters. Current-mode control monitors not only the output voltage, but also the output current. Here the voltage error signal is used to control the peak current within the magnetic elements during each power switch on-time. Current-mode control has a very rapid input and output response time, and has an inherent overcurrent protection. It is not commonly used for forward-mode converters; their current waveforms have much lower slopes in their current waveforms which can create jitter within comparators. Hysteretic control is a method of control which tries to keep a monitored parameter between two limits. There are hysteretic current and voltage control methods, but they are not commonly used. The designer should be very careful when reviewing a prospective control IC data sheet. The method of control and any variations are usually not clearly described on the first page of the data sheet.
www.onsemi.com 14
SMPSRM VCC
OSC
Charge
Clock Ramp Verror
Discharge
Ct
Volt Comp. VFB
+ -
+ +
Pulsewidth Comparator
Vref
-
Steering Average Overcurrent Protection
Cur. Comp. Iout (lavOC) or ISW (P-POC)
Output Gating Logic
Verror Amp.
Current Amp. +
RCS
+
Pulse-by-Pulse Overcurrent Protection
VOC
-
VSS
Figure 14. Voltage-Mode Control VCC
OSC
Ct
+
Discharge
Volt Comp.
Output Gating Logic S
Verror Amp.
VFB
+
Verror
Output
Q
R
+ -
Vref
Current Comparator
S
R
+ ISW
Verror
RCS VSS
Ipk ISW
Figure 15. Turn-On with Clock Current-Mode Control
www.onsemi.com 15
S
SMPSRM
The Choice of Semiconductors
One should generate a gate drive voltage that is as close to 0.7 volts as possible. This is to minimize any loss created by dropping the base drive voltage at the required base current to the level exhibited by the base. A second consideration is the storage time exhibited by the collector during its turn-off transition. When the base is overdriven, or where the base current is more than needed to sustain the collector current, the collector exhibits a 0.3-2 ms delay in its turn-off which is proportional to the base overdrive. Although the storage time is not a major source of loss, it does significantly limit the maximum switching frequency of a bipolar-based switching power supply. There are two methods of reducing the storage time and increasing its switching time. The first is to use a base speed-up capacitor whose value, typically around 100 pF, is placed in parallel with the base current limiting resistor (Figure 16a). The second is to use proportional base drive (Figure 16b). Here, only the amount of needed base current is provided by the drive circuit by bleeding the excess around the base into the collector. The last consideration with BJTs is the risk of excessive second breakdown. This phenomenon is caused by the resistance of the base across the die, permitting the furthest portions of the collector to turn off later. This forces the current being forced through the collector by an inductive load, to concentrate at the opposite ends of the die, thus causing an excessive localized heating on the die. This can result in a short-circuit failure of the BJT which can happen instantaneously if the amount of current crowding is great, or it can happen later if the amount of heating is less. Current crowding is always present when an inductive load is attached to the collector. By switching the BJT faster, with the circuits in Figure 15, one can greatly reduce the effects of second breakdown on the reliability of the device.
Power Switches The choice of which semiconductor technology to use for the power switch function is influenced by many factors such as cost, peak voltage and current, frequency of operation, and heatsinking. Each technology has its own peculiarities that must be addressed during the design phase. There are three major power switch choices: the bipolar junction transistor (BJT), the power MOSFET, and the integrated gate bipolar transistor (IGBT). The BJT was the first power switch to be used in this field and still offers many cost advantages over the others. It is also still used for very low cost or in high power switching converters. The maximum frequency of operation of bipolar transistors is less than 80-100 kHz because of some of their switching characteristics. The IGBT is used for high power switching converters, displacing many of the BJT applications. They too, though, have a slower switching characteristic which limits their frequency of operation to below 30 kHz typically although some can reach 100 kHz. IGBTs have smaller die areas than power MOSFETs of the same ratings, which typically means a lower cost. Power MOSFETs are used in the majority of applications due to their ease of use and their higher frequency capabilities. Each of the technologies will be reviewed.
The Bipolar Power Transistor The BJT is a current driven device. That means that the base current is in proportion to the current drawn through the collector. So one must provide: IB u ICńĂhFE
(eq. 8)
In power transistors, the average gain (h FE) exhibited at the higher collector currents is between 5 and 20. This could create a large base drive loss if the base drive circuit is not properly designed. VBB
VBB
+
100 pF Control IC 100 pF
Control IC
VCE + VBE -
Power Ground Power Ground (a) Fixed Base Drive Circuit
(b) Proportional Base Drive Circuit (Baker Clamp)
Figure 16. Driving a Bipolar Junction Transistor
www.onsemi.com 16
SMPSRM
The Power MOSFET
From the gate terminal, there are two capacitances the designer encounters, the gate input capacitance (Ciss) and the drain-gate reverse capacitance (Crss). The gate input capacitance is a fixed value caused by the capacitance formed between the gate metalization and the substrate. Its value usually falls in the range of 800-3200 pF, depending upon the physical construction of the MOSFET. The Crss is the capacitance between the drain and the gate, and has values in the range of 60-150 pF. Although the Crss is smaller, it has a much more pronounced effect upon the gate drive. It couples the drain voltage to the gate, thus dumping its stored charge into the gate input capacitance. The typical gate drive waveforms can be seen in Figure 18. Time period t1 is only the Ciss being charged or discharged by the impedance of the external gate drive circuit. Period t2 shows the effect of the changing drain voltage being coupled into the gate through Crss. One can readily observe the “flattening” of the gate drive voltage during this period, both during the turn-on and turn-off of the MOSFET. Time period t3 is the amount of overdrive voltage provided by the drive circuit but not really needed by the MOSFET.
Power MOSFETs are the popular choices used as power switches and synchronous rectifiers. They are, on the surface, simpler to use than BJTs, but they have some hidden complexities. A simplified model for a MOSFET can be seen in Figure 17. The capacitances seen in the model are specified within the MOSFET data sheets, but can be nonlinear and vary with their applied voltages.
CDG Coss CGS
Figure 17. The MOSFET Model
TURNON t1
VDR t3
t3
t2
t2
VGS 0
TURN-OFF
Vth
t1
Vpl
VDS 0
IG
+ 0 -
Figure 18. Typical MOSFET Drive Waveforms (Top: V GS, Middle: V DG, Bottom: IG )
www.onsemi.com 17
SMPSRM Driving MOSFETs in Switching Power Supply Applications There are three things that are very important in the high frequency driving of MOSFETs: there must be a totem-pole driver; the drive voltage source must be well bypassed; and the drive devices must be able to source high levels of current in very short periods of time (low compliance). The optimal drive circuit is shown in Figure 19.
The time needed to switch the MOSFET between on and off states is dependent upon the impedance of the gate drive circuit. It is very important that the drive circuit be bypassed with a capacitor that will keep the drive voltage constant over the drive period. A 0.1 mF capacitor is more than sufficient.
VG
VG
LOAD
LOAD
Ron
Roff
a. Passive Turn-ON
VG
b. Passive Turn-OFF
VG
LOAD
c. Bipolar Totem-pole
LOAD
d. MOS Totem-pole
Figure 19. Bipolar and FET-Based Drive Circuits (a. Bipolar Drivers, b. MOSFET Drivers)
www.onsemi.com 18
SMPSRM Sometimes it is necessary to provide a dielectrically-isolated drive to a MOSFET. This is provided by a drive transformer. Transformers driven from a DC source must be capacitively coupled from the totem-pole driver circuit. The secondary winding must be capacitively coupled to the gate with a DC restoration
T
VG
circuit. Both of the series capacitors must be more than 10 times the value of the Ciss of the MOSFET so that the capacitive voltage divider that is formed by the series capacitors does not cause an excessive attenuation. The circuit can be seen in Figure 20.
C
RG
1k C 1:1 C > 10 Ciss
Figure 20. Transformer-Isolated Gate Drive
The Insulated Gate Bipolar Transistor (IGBT)
Rectifiers Rectifiers represent about 60 percent of the losses in nonsynchronous switching power supplies. Their choice has a very large effect on the efficiency of the power supply. The significant rectifier parameters that affect the operation of switching power supplies are: • forward voltage drop (Vf ), which is the voltage across the diode when a forward current is flowing • the reverse recovery time (trr), which is how long it requires a diode to clear the minority charges from its junction area and turn off when a reverse voltage is applied • the forward recovery time (tfrr) which is how long it take a diode to begin to conduct forward current after a forward voltage is applied. There are four choices of rectifier technologies: standard, fast and ultra-fast recovery types, and Schottky barrier types. A standard recovery diode is only suitable for 50-60 Hz rectification due to its slow turn-off characteristics. These include common families such as the 1N4000 series diodes. Fast-recovery diodes were first used in switching power supplies, but their turn-off time is considered too slow for most modern applications. They may find application where low cost is paramount, however. Ultra-fast recovery diodes turn off quickly and have a forward voltage drop of 0.8 to 1.3 V, together with a high reverse voltage capability of up to 1000 V. A Schottky rectifier turns off very quickly and has an average forward voltage drop of between 0.35 and 0.8 V, but has a low reverse breakdown voltage and
The IGBT is a hybrid device with a MOSFET as the input device, which then drives a silicon-controlled rectifier (SCR) as a switched output device. The SCR is constructed such that it does not exhibit the latching characteristic of a typical SCR by making its feedback gain less than 1. The die area of the typical IGBT is less than one-half that of an identically rated power MOSFET, which makes it less expensive for high-power converters. The only drawback is the turn-off characteristic of the IGBT. Being a bipolar minority carrier device, charges must be removed from the P-N junctions during a turn-off condition. This causes a “current tail” at the end of the turn-off transition of the current waveform. This can be a significant loss because the voltage across the IGBT is very high at that moment. This makes the IGBT useful only for frequencies typically less than 20 kHz, or for exceptional IGBTs, 100 kHz. To drive an IGBT one uses the MOSFET drive circuits shown in Figures 18 and 19. Driving the IGBT gate faster makes very little difference in the performance of an IGBT, so some reduction in drive currents can be used. The voltage drop of across the collector-to-emitter (VCE) terminals is comparable to those found in Darlington BJTs and MOSFETs operated at high currents. The typical VCE of an IGBT is a flat 1.5-2.2 volts. MOSFETs, acting more resistive, can have voltage drops of up to 5 volts at the end of some high current ramps. This makes the IGBT, in high current environments, very comparable to MOSFETs in applications of less than 5-30 kHz.
www.onsemi.com 19
SMPSRM a high reverse leakage current. For a typical switching power supply application, the best choice is usually a Schottky rectifier for output voltages less than 12 V, and an ultra-fast recovery diode for all other output voltages. The major losses within output rectifiers are conduction losses and switching losses. The conduction loss is the forward voltage drop times the current flowing through it during its conduction period. This can be significant if its voltage drop and current are high. The switching losses are determined by how fast a diode turns off (trr) times the reverse voltage across the rectifier. This can be significant for high output voltages and currents.
The characteristics of power rectifiers and their applications in switching power supplies are covered in great detail in Reference (5). The major losses within output rectifiers are conduction losses and switching losses. The conduction loss is the forward voltage drop times the current flowing through it during its conduction period. This can be significant if its voltage drop and current are high. The switching losses are determined by how fast a diode turns off (trr) times the reverse voltage across the rectifier. This can be significant for high output voltages and currents.
Table 2. Types of Rectifier Technologies Rectifier Type
Average Vf
Reverse Recovery Time
Typical Applications
Standard Recovery
0.7-1.0 V
1,000 ns
50-60 Hz Rectification
Fast Recovery
1.0-1.2 V
150-200 ns
Output Rectification
UltraFast Recovery
0.9-1.4 V
25-75 ns
Output Rectification (Vo > 12 V)
Schottky
0.3-0.8 V
< 10 ns
Output Rectification (Vo < 12 V)
Table 3. Estimating the Significant Parameters of the Power Semiconductors Topology
Bipolar Pwr Sw
MOSFET Pwr Sw
Rectifier
V CEO
IC
V DSS
ID
V R
IF
Buck
Vin
Iout
Vin
Iout
Vin
Iout
Boost
Vout
(2.0ĂPout) Vin(min)
Vout
(2.0ĂPout) Vin(min)
Vout
Iout
Buck/Boost
Vin * Vout
(2.0ĂPout) Vin(min)
Vin * Vout
(2.0ĂPout) Vin(min)
Vin * Vout
Iout
1.7ĂVin(max)
(2.0ĂPout) Vin(min)
1.5ĂVin(max)
(2.0ĂPout) Vin(min)
5.0 Vout
Iout
Flyback 1 Transistor Forward
2.0 Vin
(1.5ĂPout) Vin(min)
2.0 Vin
(1.5ĂPout) Vin(min)
3.0 Vout
Iout
Push-Pull
2.0 Vin
(1.2ĂPout) Vin(min)
2.0 Vin
(1.2ĂPout) Vin(min)
2.0 Vout
Iout
Half-Bridge
Vin
(2.0ĂPout) Vin(min)
Vin
(2.0ĂPout) Vin(min)
2.0 Vout
Iout
Full-Bridge
Vin
(1.2ĂPout) Vin(min)
Vin
(2.0ĂPout) Vin(min)
2.0 Vout
Iout
www.onsemi.com 20
SMPSRM
The Magnetic Components
Coiltronics, Division of Cooper Electronics Technology
The magnetic elements within a switching power supply are used either for stepping-up or down a switched AC voltage, or for energy storage. In forward-mode topologies, the transformer is only used for stepping-up or down the AC voltage generated by the power switches. The output filter (the output inductor and capacitor) in forward-mode topologies is used for energy storage. In boost-mode topologies, the transformer is used both for energy storage and to provide a step-up or step-down function. Many design engineers consider the magnetic elements of switching power supplies counter-intuitive or too complicated to design. Fortunately, help is at hand; the suppliers of magnetic components have applications engineers who are quite capable of performing the transformer design and discussing the tradeoffs needed for success. For those who are more experienced or more adventuresome, please refer to Reference 2 in the Bibliography for transformer design guidelines. The general procedure in the design of any magnetic component is as follows (Reference 2, p 42): 1. Select an appropriate core material for the application and the frequency of operation. 2. Select a core form factor that is appropriate for the application and that satisfies applicable regulatory requirements. 3. Determine the core cross-sectional area necessary to handle the required power 4. Determine whether an airgap is needed and calculate the number of turns needed for each winding. Then determine whether the accuracy of the output voltages meets the requirements and whether the windings will fit into the selected core size. 5. Wind the magnetic component using proper winding techniques. 6. During the prototype stage, verify the component's operation with respect to the level of voltage spikes, cross-regulation, output accuracy and ripple, RFI, etc., and make corrections were necessary. The design of any magnetic component is a “calculated estimate.” There are methods of “stretching” the design limits for smaller size or lower losses, but these tend to be diametrically opposed to one another. One should be cautious when doing this. Some useful sources for magnetics components are:
6000 Park of Commerce Blvd Boca Raton, FL (USA) 33487 website: http://www.coiltronics.com Telephone: 561-241-7876 Cramer Coil, Inc.
401 Progress Dr. Saukville, WI (USA) 53080 website: http://www.cramerco.com email:
[email protected] Telephone: 262-268-2150 Pulse, Inc.
San Diego, CA website: http://www.pulseeng.com Telephone: 858-674-8100 TDK
1600 Feehanville Drive Mount Prospect, IL 60056 website: http://www.component.talk.com Telephone: 847-803-6100
Laying Out the Printed Circuit Board The printed circuit board (PCB) layout is the third critical portion of every switching power supply design in addition to the basic design and the magnetics design. Improper layout can adversely affect RFI radiation, component reliability, efficiency and stability. Every PCB layout will be different, but if the designer appreciates the common factors present in all switching power supplies, the process will be simplified. All PCB traces exhibit inductance and resistance. These can cause high voltage transitions whenever there is a high rate of change in current flowing through the trace. For operational amplifiers sharing a trace with power signals, it means that the supply would be impossible to stabilize. For traces that are too narrow for the current flowing through them, it means a voltage drop from one end of the trace to the other which potentially can be an antenna for RFI. In addition, capacitive coupling between adjacent traces can interfere with proper circuit operation. There are two rules of thumb for PCB layouts: “short and fat” for all power-carrying traces and “one point grounding” for the various ground systems within a switching power supply. Traces that are short and fat minimize the inductive and resistive aspects of the trace, thus reducing noise within the circuits and RFI. Single-point grounding keeps the noise sources separated from the sensitive control circuits.
CoilCraft, Inc.
1102 Silver Lake Rd. Cary, IL (USA) 60013 website: http://www.coilcraft.com/ email:
[email protected] Telephone: 847-639-6400
www.onsemi.com 21
SMPSRM Within all switching power supplies, there are four major current loops. Two of the loops conduct the high-level AC currents needed by the supply. These are the power switch AC current loop and the output rectifier AC current loop. The currents are the typical trapezoidal current pulses with very high peak currents and very rapid di/dts. The other two current loops are the input source and the output load current loops, which carry low frequency current being supplied from the voltage source and to the load respectively. For the power switch AC current loop, current flows from the input filter capacitor through the inductor or transformer winding, through the power switch and back to the negative pin of the input capacitor. Similarly, the output rectifier current loop's current flows from the inductor or secondary transformer winding, through the
rectifier to the output filter capacitor and back to the inductor or winding. The filter capacitors are the only components that can source and sink the large levels of AC current in the time needed by the switching power supply. The PCB traces should be made as wide and as short as possible, to minimize resistive and inductive effects. These traces should be the first to be laid out. Turning to the input source and output load current loops, both of these loops must be connected directly to their respective filter capacitor's terminals, otherwise switching noise could bypass the filtering action of the capacitor and escape into the environment. This noise is called conducted interference. These loops can be seen in Figure 21 for the two major forms of switching power supplies, non-isolated (Figure 21a) and transformer-isolated (Figure 21b).
ower w tc Current Loop
utput ect er Current Loop
L
Vout
SW
Input Current Loop
Output Load Current Loop + Vin -
VFB
Control
Cin
Cout
Analog
GND C
A Input Source Ground
B Power Switch Ground
Join
Output Load Ground
Output Rectifier Ground Join
Join (a) The Non-Isolated DC/DC Converter
Input Current Loop
Output Rectifier Current Loop
Power Switch Current Loop
Output Load Current Loop Vout
VFB SW
+ Vin -
Cout
Cin
Control
FB
RCS
Analog GND
C A Input Source Ground
B Output Rectifier Ground
Output Load Ground Join
Join
Power Switch Ground Join
(b) The Transformer-Isolated Converter
Figure 21. The Current Loops and Grounds for the Major Converter Topologies
www.onsemi.com 22
SMPSRM The grounds are extremely important to the proper operation of the switching power supply, since they form the reference connections for the entire supply; each ground has its own unique set of signals which can adversely affect the operation of the supply if connected improperly. There are five distinct grounds within the typical switching power supply. Four of them form the return paths for the current loops described above. The remaining ground is the low-level analog control ground which is critical for the proper operation of the supply. The grounds which are part of the major current loops must be connected together exactly as shown in Figure 21. Here again, the connecting point between the high-level AC grounds and the input or output grounds is at the negative terminal of the appropriate filter capacitor (points A and B in Figures 21a and 21b). Noise on the AC grounds can very easily escape into the environment if the grounds are not directly connected to the negative terminal of the filter capacitor(s). The analog control ground must be connected to the point where the control IC and associated circuitry must measure key power parameters, such as AC or DC current and the output voltage (point C in Figures 21a and 21b). Here any noise introduced by large AC signals within the AC grounds will sum directly onto the low-level control parameters and greatly affect the operation of the supply. The purpose of connecting the control ground to the lower side of the current sensing resistor or the output voltage resistor divider is to form a “Kelvin contact” where any common mode noise is not sensed by the control circuit. In short, follow the example given by Figure 21 exactly as shown for best results.
The last important factor in the PCB design is the layout surrounding the AC voltage nodes. These are the drain of the power MOSFET (or collector of a BJT) and the anode of the output rectifier(s). These nodes can capacitively couple into any trace on different layers of the PCB that run underneath the AC pad. In surface mount designs, these nodes also need to be large enough to provide heatsinking for the power switch or rectifier. This is at odds with the desire to keep the pad as small as possible to discourage capacitive coupling to other traces. One good compromise is to make all layers below the AC node identical to the AC node and connect them with many vias (plated-through holes). This greatly increases the thermal mass of the pad for improved heatsinking and locates any surrounding traces off laterally where the coupling capacitance is much smaller. An example of this can be seen in Figure 22. Many times it is necessary to parallel filter capacitors to reduce the amount of RMS ripple current each capacitor experiences. Close attention should be paid to this layout. If the paralleled capacitors are in a line, the capacitor closest to the source of the ripple current will operate hotter than the others, shortening its operating life; the others will not see this level of AC current. To ensure that they will evenly share the ripple current, ideally, any paralleled capacitors should be laid out in a radially-symmetric manner around the current source, typically a rectifier or power switch. The PCB layout, if not done properly, can ruin a good paper design. It is important to follow these basic guidelines and monitor the layout every step of the process.
Power Device
Via
Plated-Thru Hole
PCB Top
PCB Bottom
Figure 22. Method for Minimizing AC Capacitive Coupling and Enhancing Heatsinking
www.onsemi.com 23
SMPSRM
Losses and Stresses in Switching Power Supplies
the circuitry, and some are controlled by simply selecting a different part. Identifying the major sources for loss can be as easy as placing a finger on each of the components in search of heat, or measuring the currents and voltages associated with each power component using an oscilloscope, AC current probe and voltage probe. Semiconductor losses fall into two categories: conduction losses and switching losses. The conduction loss is the product of the terminal voltage and current during the power device's on period. Examples of conduction losses are the saturation voltage of a bipolar power transistor and the “on” loss of a power MOSFET shown in Figure 23 and Figure 24 respectively.
Much of the designer's time during a switching power supply design is spent in identifying and minimizing the losses within the supply. Most of the losses occur in the power components within the switching power supply. Some of these losses can also present stresses to the power semiconductors which may affect the long term reliability of the power supply, so knowing where they arise and how to control them is important. Whenever there is a simultaneous voltage drop across a component with a current flowing through, there is a loss. Some of these losses are controllable by modifying
R E T T I M E ‐ ) S O T T ‐ L R O V O T ( C E L L O C
E G A T L O V E ) S C T R L U O O V S ‐ ( O T ‐ N I A R D RISE TIME
VPEAK
SATURATION VOLTAGE
RISE DYNAMIC TIME SATURATION
FALL TIME
STORAGE TIME
VPEAK
ON VOLTAGE FALL TIME
CLEARING RECTIFIERS
CLEARING T IPEAK RECTIFIERS N E R R U ) PINCHING OFF INDUCTIVE S C P R CHARACTERISTICS OF THE M O TRANSFORMER T A ( C E SATURATION L L CURRENT O TURN‐ON TURN‐OFF CURRENT C CURRENT CURRENT TAIL
T N E R ) R S U P C M ( N A I A R D
PINCHING OFF INDUCTIVE CHARACTERISTICS OF THE TRANSFORMER ON CURRENT TURN‐ON CURRENT
CURRENT CROWDING PERIOD
Y G R E ) N S E E S L U U O O J E ( N S A S T O N L A T S TURN‐ON N I LOSS
SECOND Y BREAKDOWN G R PERIOD E ) N S E E S L U U O O ( E J N S A S SATURATION T O N LOSS L A T TURN‐ON TURN‐OFF LOSS S LOSS SWITCHING LOSS N I
Figure 23. Stresses and Losses within a Bipolar Power Transistor
IPEAK
TURN‐OFF CURRENT
ON LOSS TURN‐OFF LOSS SWITCHING LOSS
Figure 24. Stresses and Losses within a Power MOSFET
www.onsemi.com 24
SMPSRM The forward conduction loss of a rectifier is shown in Figure 25. During turn-off, the rectifier exhibits a reverse recovery loss where minority carriers trapped within the P-N junction must reverse their direction and exit the junction after a reverse voltage is applied. This results in what appears to be a current flowing in reverse through the diode with a high reverse terminal voltage. The switching loss is the instantaneous product of the terminal voltage and current of a power device when it is transitioning between operating states (on-to-off and off-to-on). Here, voltages are transitional between full-on and cutoff states while simultaneously the current is transitional between full-on and cut-off states. This E G A ) T L S O T V L O E ( V D O I D
creates a very large V-I product which is as significant as the conduction losses. Switching losses are also the major frequency dependent loss within every PWM switching power supply. The loss-induced heat generation causes stress within the power component. This can be minimized by an effective thermal design. For bipolar power transistors, however, excessive switching losses can also provide a lethal stress to the transistor in the form of second breakdown and current crowding failures. Care should be taken in the careful analysis of each transistor's Forward Biased-Safe Operating Area (FBSOA) and Reverse Biased-Safe Operating Area (RBSOA) operation.
FORWARD VOLTAGE
REVERSE VOLTAGE
IPK T N E R ) R S U P C M E A ( D O I D
FORWARD CONDUCTION CURRENT DEGREE OF DIODE RECOVERY ABRUPTNESS
FORWARD RECOVERY TIME (Tfr) Y G R E ) N S E E S L U U O O ( E J N S A S T O N L A T S N I
REVERSE RECOVERY TIME (Trr)
FORWARD CONDUCTION LOSS SWITCHING LOSS
Figure 25. Stresses and Losses within Rectifiers
Techniques to Improve Efficiency in Switching Power Supplies
rectification is a technique to reduce this conduction loss by using a switch in place of the diode. The synchronous rectifier switch is open when the power switch is closed, and closed when the power switch is open, and is typically a MOSFET inserted in place of the output rectifier. To prevent ”crowbar” current that would flow if both switches were closed at the same time, the switching scheme must be break-before-make. Because of this, a diode is still required to conduct the initial current during the interval between the opening of the main switch and the closing of the synchronous rectifier switch. A Schottky rectifier with a current rating of 30 percent of
The reduction of losses is important to the efficient operation of a switching power supply, and a great deal of time is spent during the design phase to minimize these losses. Some common techniques are described below. The Synchronous Rectifier As output voltages decrease, the losses due to the output rectifier become increasingly significant. For Vout = 3.3 V, a typical Schottky diode forward voltage of 0.4 V leads to a 12% loss of efficiency. Synchronous
www.onsemi.com 25
SMPSRM the MOSFET should be placed in parallel with the synchronous MOSFET. The MOSFET does contain a parasitic body diode that could conduct current, but it is lossy, slow to turn off, and can lower efficiency by 1% to 2%. The lower turn-on voltage of the Schottky prevents the parasitic diode from ever conducting and exhibiting its poor reverse recovery characteristic. Using synchronous rectification, the conduction voltage can be reduced from 400 mV to 100 mV or less. An improvement of 1-5 percent can be expected for the
typical switching power supply. The synchronous rectifier can be driven either actively, that is directly controlled from the control IC, or passively, driven from other signals within the power circuit. It is very important to provide a non-overlapping drive between the power switch(es) and the synchronous rectifier(s) to prevent any shoot-through currents. This dead time is usually between 50 to 100 ns. Some typical circuits can be seen in Figure 26.
Vin
+ Vout
-
SW Drive GND Direct
SR
RG
C VG C
D
1k
1:1 C > 10 C iss Transformer-Isolated (a) Actively Driven Synchronous Rectifiers
LO + Vout Primary -
(b) Passively Driven Synchronous Rectifiers
Figure 26. Synchronous Rectifier Circuits
www.onsemi.com 26
SMPSRM Snubbers and Clamps Snubbers and clamps are used for two very different purposes. When misapplied, the reliability of the semiconductors within the power supply is greatly jeopardized. A snubber is used to reduce the level of a voltage spike and decrease the rate of change of a voltage waveform. This then reduces the amount of overlap of the voltage and current waveforms during a transition, thus reducing the switching loss. This has its benefits in the Safe Operating Area (SOA) of the semiconductors, and it reduces emissions by lowering the spectral content of any RFI. A clamp is used only for reducing the level of a voltage spike. It has no affect on the dV/dt of the transition.
ZENER CLAMP
SOFT CLAMP
Therefore it is not very useful for reducing RFI. It is useful for preventing components such as semiconductors and capacitors from entering avalanche breakdown. Bipolar power transistors suffer from current crowding which is an instantaneous failure mode. If a voltage spike occurs during the turn-off voltage transition of greater than 75 percent of its VCEO rating, it may have too much current crowding stress. Here both the rate of change of the voltage and the peak voltage of the spike must be controlled. A snubber is needed to bring the transistor within its RBSOA (Reverse Bias Safe Operating Area) rating. Typical snubber and clamp circuits are shown in Figure 27. The effects that these have on a representative switching waveform are shown in Figure 28.
SNUBBER
SNUBBER
SOFT CLAMP
ZENER CLAMP
Figure 27. Common Methods for Controlling Voltage Spikes and/or RFI
CLAMP ) S T L O V ( E G A T L O V
SNUBBER
ORIGINAL WAVEFORM
t, TIME ( μsec)
Figure 28. The Effects of a Snubber versus a Clamp
www.onsemi.com 27
SMPSRM The Lossless Snubber A lossless snubber is a snubber whose trapped energy is recovered by the power circuit. The lossless snubber is designed to absorb a fixed amount of energy from the transition of a switched AC voltage node. This energy is stored in a capacitor whose size dictates how much energy the snubber can absorb. A typical implementation of a lossless snubber can be seen in Figure 29. The design for a lossless snubber varies from topology to topology and for each desired transition. Some adaptation may be necessary for each circuit. The important factors in the design of a lossless snubber are: 1. The snubber must have initial conditions that allow it to operate during the desired transition and at the desired voltages. Lossless snubbers should be emptied of their energy prior to the desired transition. The voltage to which it is reset dictates where the snubber will begin to operate. So if the snubber is reset to the input voltage, then it will act as a lossless clamp which will remove any spikes above the input voltage.
2. When the lossless snubber is “reset,” the energy should be returned to the input capacitor or back into the output power path. Study the supply carefully. Returning the energy to the input capacitor allows the supply to use the energy again on the next cycle. Returning the energy to ground in a boostmode supply does not return the energy for reuse, but acts as a shunt current path around the power switch. Sometimes additional transformer windings are used. 3. The reset current waveform should be band limited with a series inductor to prevent additional EMI from being generated. Use of a 2 to 3 turn spiral PCB inductor is sufficient to greatly lower the di/dt of the energy exiting the lossless snubber.
Unsnubbed VSW
+ VSW
Snubbed VSW ID
-
Drain Current (ID)
Figure 29. Lossless Snubber for a One Transistor Forward or Flyback Converter
www.onsemi.com 28
SMPSRM The Active Clamp An active clamp is a gated MOSFET circuit that allows the controller IC to activate a clamp or a snubber circuit at a particular moment in a switching power supply's cycle of operation. An active clamp for a flyback converter is shown in Figure 30. In Figure 30, the active clamp is reset (or emptied of its
stored energy) just prior to the turn-off transition. It is then disabled during the negative transition. Obviously, the implementation of an active clamp is more expensive than other approaches, and is usually reserved for very compact power supplies where heat is a critical issue.
Unclamped Switch Voltage (VSW)
Clamped Switch Voltage (VSW) Vin Switch Current (ISW)
+ ICL
VDR
+
ISW
VSW -
GND
Drive Voltage (VDR) Discharge
Charge
Clamp Current (ICL)
Figure 30. An Active Clamp Used in a One Transistor Forward or a Flyback Converter
www.onsemi.com 29
SMPSRM Quasi-Resonant Topologies A quasi-resonant topology is designed to reduce or eliminate the frequency-dependent switching losses within the power switches and rectifiers. Switching losses account for about 40% of the total loss within a PWM power supply and are proportional to the switching frequency. Eliminating these losses allows the designer to increase the operating frequency of the switching power supply and so use smaller inductors and capacitors, reducing size and weight. In addition, RFI levels are reduced due to the controlled rate of change of current or voltage. The downside to quasi-resonant designs is that they are more complex than non-resonant topologies due to parasitic RF effects that must be considered when
switching frequencies are in the 100's of kHz. Schematically, quasi-resonant topologies are minor modifications of the standard PWM topologies. A resonant tank circuit is added to the power switch section to make either the current or the voltage “ring” through a half a sinusoid waveform. Since the sinusoid starts at zero and ends at zero, the product of the voltage and current at the starting and ending points is zero, thus has no switching loss. There are two quasi-resonant methods: zero current switching (ZCS) or zero voltage switching (ZVS). ZCS is a fixed on-time, variable off-time method of control. ZCS starts from an initial condition where the power switch is off and no current is flowing through the resonant inductor. The ZCS quasi-resonant buck converter is shown in Figure 31.
ILR LR
CR
VSW
Vin Cin
LO
D Cout
CONTROL
Vout
FEEDBACK
A ZCS Quasi-Resonant Buck Converter SWITCH TURN‐OFF
W S V
Vin POWER SWITCH ON
IPK R L I
D V
Figure 31. Schematic and Waveforms for a ZCS Quasi‐Resonant Buck Converter
www.onsemi.com 30
SMPSRM In this design, both the power switch and the catch diode operate in a zero current switching mode. Power is passed to the output during the resonant periods. So to increase the power delivered to the load, the frequency would increase, and vice versa for decreasing loads. In typical designs the frequency can change 10:1 over the ZCS supply's operating range. The ZVS is a fixed off-time, variable on-time method control. Here the initial condition occurs when the power switch is on, and the familiar current ramp is flowing through the filter inductor. The ZVS quasi-resonant buck converter is shown in Figure 32. Here, to control the
power delivered to the load, the amount of “resonant off times” are varied. For light loads, the frequency is high. When the load is heavy, the frequency drops. In a typical ZVS power supply, the frequency typically varies 4:1 over the entire operating range of the supply. There are other variations on the resonant theme that promote zero switching losses, such as full resonant PWM, full and half-bridge topologies for higher power and resonant transition topologies. For a more detailed treatment, see Chapter 4 in the “Power Supply Cookbook” (Bibliography reference 2).
LR
LO
CR Vin
Cin
D
VI/P
FEEDBACK
CONTROL
A ZVS Quasi-Resonant Buck Converter
Vin P / I V
POWER SWITCH TURNS ON
0
V IPK W S I
in
L V L
*V
R
out
)L
O
in R
0
ILOAD
D I
Figure 32. Schematic and Waveforms for a ZVS Quasi‐Resonant Buck Converter
www.onsemi.com 31
Cout
Vout
SMPSRM
Power Factor Correction
requiring all electrical equipment connected to a low voltage distribution system to minimize current harmonics and maximize power factor. 2. The reflected power not wasted in the resistance of the power cord may generate unnecessary heat in the source (the local step-down transformer), contributing to premature failure and constituting a fire hazard. 3. Since the ac mains are limited to a finite current by their circuit breakers, it is desirable to get the most power possible from the given current available. This can only happen when the power factor is close to or equal to unity. The typical AC input rectification circuit is a diode bridge followed by a large input filter capacitor. During the time that the bridge diodes conduct, the AC line is driving an electrolytic capacitor, a nearly reactive load. This circuit will only draw current from the input lines when the input's voltage exceeds the voltage of the filter capacitor. This leads to very high currents near the peaks of the input AC voltage waveform as seen in Figure 33. Since the conduction periods of the rectifiers are small, the peak value of the current can be 3-5 times the average input current needed by the equipment. A circuit breaker only senses average current, so it will not trip when the peak current becomes unsafe, as found in many office areas. This can present a fire hazard. In three-phase distribution systems, these current peaks sum onto the neutral line, not meant to carry this kind of current, which again presents a fire hazard.
Power Factor (PF) is defined as the ratio of real power to apparent power. In a typical AC power supply application where both the voltage and current are sinusoidal, the PF is given by the cosine of the phase angle between the input current and the input voltage and is a measure of how much of the current contributes to real power in the load. A power factor of unity indicates that 100% of the current is contributing to power in the load while a power factor of zero indicates that none of the current contributes to power in the load. Purely resistive loads have a power factor of unity; the current through them is directly proportional to the applied voltage. The current in an ac line can be thought of as consisting of two components: real and imaginary. The real part results in power absorbed by the load while the imaginary part is power being reflected back into the source, such as is the case when current and voltage are of opposite polarity and their product, power, is negative. It is important to have a power factor as close as possible to unity so that none of the delivered power is reflected back to the source. Reflected power is undesirable for three reasons: 1. The transmission lines or power cord will generate heat according to the total current being carried, the real part plus the reflected part. This causes problems for the electric utilities and has prompted various regulations
Power not used E G A T L O V
Power used 110/220 AC VOLTS IN
I + Clarge
DC To Power Supply -
T N E R R U C
I AV
Figure 33. The Waveforms of a Capacitive Input Filter
www.onsemi.com 32
SMPSRM A Power Factor Correction (PFC) circuit is a switching power converter, essentially a boost converter with a very wide input range, that precisely controls its input current on an instantaneous basis to match the waveshape and phase of the input voltage. This represents a zero degrees or 100 percent power factor and mimics a purely resistive load. The amplitude of the input current waveform is varied over longer time frames to maintain a constant voltage at the converter's output filter capacitor. This mimics a resistor which slowly changes value to absorb the correct amount of power to meet the demand of the load. Short term energy excesses and deficits caused by sudden changes in the load are supplemented by a ”bulk energy storage capacitor”, the boost converter's output filter device. The PFC input filter capacitor is reduced to a few microfarads, thus placing a half-wave haversine waveshape into the PFC converter. The PFC boost converter can operate down to about 30 V before there is insufficient voltage to draw any more significant power from its input. The converter then can begin again when the input haversine reaches 30 V on the next half-wave haversine. This greatly increases the conduction angle of the input rectifiers. The drop-out region of the PFC converter is then filtered (smoothed) by the input EMI filter. A PFC circuit not only ensures that no power is reflected back to the source, it also eliminates the high current pulses associated with conventional rectifier-filter input circuits. Because heat lost in the transmission line and adjacent circuits is proportional to the square of the current in the line, short strong current
pulses generate more heat than a purely resistive load of the same power. The active power factor correction circuit is placed just following the AC rectifier bridge. An example can be seen in Figure 34. Depending upon how much power is drawn by the unit, there is a choice of three different common control modes. All of the schematics for the power sections are the same, but the value of the PFC inductor and the control method are different. For input currents of less than 150 watts, a discontinuous-mode control scheme is typically used, in which the PFC core is completely emptied prior to the next power switch conduction cycle. For powers between 150 and 250 watts, the critical conduction mode is recommended. This is a method of control where the control IC senses just when the PFC core is emptied of its energy and the next power switch conduction cycle is immediately begun; this eliminates any dead time exhibited in the discontinuous-mode of control. For an input power greater than 250 watts, the continuous-mode of control is recommended. Here the peak currents can be lowered by the use of a larger inductor, but a troublesome reverse recovery characteristic of the output rectifier is encountered, which can add an additional 20-40 percent in losses to the PFC circuit. Many countries cooperate in the coordination of their power factor requirements. The most appropriate document is IEC61000-3-2, which encompasses the performance of generalized electronic products. There are more detailed specifications for particular products made for special markets.
Switch Current Input Voltage
I
Vout Vsense
Csmall
+ Control
Conduction Angle
Figure 34. Power Factor Correction Circuit e g a t l o V
I AVG
t n e r r u C
Figure 35. Waveform of Corrected Input
www.onsemi.com 33
Clarge
To Power Supply -
SMPSRM
Bibliography 1. Ben-Yaakov Sam, Gregory Ivensky, “Passive Lossless Snubbers for High Frequency PWM Converters,” Seminar 12, APEC 99. 2. Brown, Marty, Power Supply Cookbook , Butterworth-Heinemann, 1994, 2001. 3. Brown, Marty, “Laying Out PC Boards for Embedded Switching Supplies,” Electronic Design, Dec. 1999. 4. Martin, Robert F., “Harmonic Currents,” Compliance Engineering - 1999 Annual Resources Guide , Cannon Communications, LLC, pp. 103-107. 5. ON Semiconductor, Rectifier Applications Handbook , HB214/D, Rev. 2, Nov. 2001.
www.onsemi.com 34
SMPSRM
SWITCHMODE Power Supply Examples This section provides both initial and detailed information to simplify the selection and design of a variety of SWITCHMODE power supplies. The ICs for Switching Power Supplies figure identifies control, reference voltage, output protection and switching regulator ICs for various topologies. Page ICs for Switching Power Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Integrated circuits identified for various sections of a switching power supply.
36
Suggested Components for Specific Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37 A list of suggested control ICs, power transistors and rectifiers for SWITCHMODE power supplies by application. •ăCRT Display System . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . •ăAC/DC Power Supply for CRT Displays . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . •ăAC/DC Power Supply for Storage, Imaging & Entertainment . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . •ăDC-DC Conversion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . •ăTypical PC Forward-Mode SMPS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
38 39 39 40 41
Real SMPS Applications 80 W Power Factor Correction Controller . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Compact Power Factor Correction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Monitor Pulsed-Mode SMPS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 70 W Wide Mains TV SMPS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 100 W Wide Mains TV SMPS with 1.3 W Stand-by . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Low-Cost Off-line IGBT Battery Charger . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 110 W Output Flyback SMPS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Efficient Safety Circuit for Electronic Ballast . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . AC-DC Battery Charger - Constant Current with Voltage Limit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
42 43 44 46 48 50 51 53 55
Some of these circuits may have a more complete application note, spice model information or even an evaluation board available. Consult ON Semiconductor's website ( www.onsemi.com ) or local sales office for more information.
www.onsemi.com 35
SMPSRM
Figure 36. . Intergrated Ciruits for Switching Power Supplies
N O I S R E V N O C C D C D
S R E T L I F T U P T U O
/ R E P B M B A U L N C S
A 3 6 7 A 2 0 1 7 A 0 3 6 6 6 0 0 1 1 1 0 5 6 1 1 1 0 4 4 4 4 5 5 0 4 4 4 4 1 1 1 1 4 1 4 3 3 3 1 P P P P 1 P 3 C C C P P C C C C C C C C M M M M N N N N N N N 6 1 3 1 2 3 4 1 2 3 4 3 6 3 3 1 1 1 1 7 7 7 7 6 4 0 0 4 4 4 4 1 1 1 1 4 3 1 1 1 1 1 1 5 5 5 5 3 5 5 5 5 5 5 S S S S 3 3 S S S S S S C C C C C C C C C C C C M M
T L 0 C 0 0 6 0 6 4 3 6 3 2 S 8 S S R F R R B H U B R M M U M M
N O I T C E T O R P T U P T U O
1 4 3 x 3 6 6 2 0 0 1 1 4 3 8 3 3 3 P P 3 3 C C C C C M M M N N 7 8 9 0 4 0 0 0 1 6 7 7 8 8 0 X X X X 3 A A A A 3 C M M M M M
R O N T I O C T A C F E R R E R W O O C P
N O I C S D R - E C V D N O C
N O T I U T P C T E U T O O R P E K C G A A B T L D O E V E F
E K C G A A B T D L O E V E F
7 4 3 5 5 7 4 A 2 2 2 2 6 6 6 3 2 1 0 0 0 0 3 0 1 5 3 3 3 3 3 6 5 S 3 3 3 3 3 4 S C C C C C 4 C C C M M M M M M
/ R E P B M B A U L N C S
F E R M W P
S O S M R R E E V I R W O D P
0 2 8 2 0 1 6 6 6 6 5 5 2 2 3 2 6 6 3 3 3 4 1 1 3 3 3 3 P P C C C C C C M M M M N N
B / A 1 0 1 0 0 A 3 1 1 / 4 5 P 1 3 S C 4 V C N L L T T
4 5 8 0 5 x 0 0 0 0 0 4 6 6 6 2 2 8 4 4 4 1 1 3 4 4 4 P P C C C C C C M M M N N U
- S S R N E A M R R T O F
0 A A A A 0 0 0 6 x x x x x 6 6 6 2 x x x 1 2 1 x 1 B 2 3 R R S S E 6 6 U U R R K M N N U U 6 S M 1 1 M M M P 6 P
1 2 3 5 5 5 1 1 1 3 3 3 3 3 3 C C C M M M
A 6 8 A 2 5 8 4 x 7 x x 5 0 0 8 1 1 1 2 x 1 0 1 0 9 9 0 9 9 9 9 M 1 5 9 7 7 9 7 7 7 7 9 1 5 7 C C 7 C C C C 7 P P P C C C C C C N M M M M M M M M M N N N O I T A L x x 2 x x x 5 1 8 x x . U x 5 x x x x x 7 6 6 x 5 0 C 2 7 2 8 B F L M P T 9 0 G 3 3 4 7 8 8 8 8 8 8 7 9 E 3 3 3 C 7 7 7 7 7 7 C 7 R C C C C C C C C C M C E M M M M M M M M M M M G A T L 3 9 O 1 5 L 0 1 M 6 6 7 0 9 3 3 7 7 7 3 5 5 5 2 2 V 4 9 9 1 1 3 3 9 9 3 3 9 2 2 3 1 3 M M 2 2 3 3 4 M M M 3 P P C C L L L L M L M L L L L L M M
f e r V
T S U R P E T T U L I O F
0 0 0 0 0 0 0 0 6 6 1 3 3 1 1 1 3 R D 1 R R B R S B B B R B M M M M M
S O S M R R E V E I R W O D P
N O E I G T A A L T L U O G V E R
C S O
L O R T N O C
R H E C T I W O W P S P U T R A T S
R O N T I O C T A C F E R R E R W O O C P
2 3 4 6 0 1 3 1 2 2 2 0 2 2 4 2 0 0 0 1 2 2 8 0 1 1 1 5 1 1 3 1 5 5 5 S 5 5 S S 5 S S S S C C S C C C C C C
x x x x x x 2 2 6 5 5 4 Z Z Z B S S M M M M M M
2 3 5 x x 6 6 6 0 5 3 3 3 0 0 3 3 3 1 1 3 3 3 P P C C C C C M M M N N
www.onsemi.com 36
G S N R I H O C T T A I L U W S G V E H R
L O R T N O C
P U T R A T S
s e i l p p u S r e w o P g n i h c t i w S r o f s t i u c r i C d e t a r g e t n I . 0 3 e r u g i F
SMPSRM e d E E o 0 0 0 i 0 6 D 0 1 4 r 1 4 R e 8 p R R U m U U M a M M D
T R C
R 5 / T 1 r 0 e 1 v N i r 6 D D - T H M
B G R
R
M W R
G B G R
c n y S _ V
c n y S _ H
1 0 1 0 0 1 1 0 1 0 1
r o U t i n C o M M
R
B C G M 2 W I R P r o
Y R O M E M
R O S S E C O R P C N Y S
E 5 U 0 R P C O C H C 0 4 8 x 2 2 0 1 1
c n y S _ V
c n y S _ H
B
B U H B S U
N W O D
0 4 8 / 0 3 8 / 0 4 7 / 0 3 7 / 0 4 6 / 0 3 6 F R I
n o i t c e r r o C y r t e m o e G
B / x x 0 A 0 5 0 3 0 0 2 6 1 1 1 3 P P P 3 3 C C C C N N N M
P U
3 E / 2 0 4 2 8 P 3 6 C P T U M
R E C L D L O O R T T C N D O C
B
G
d e y B a G l Figure 37. r e R v . 15” O Monitor Power Supplies y a l p r s o i t D r n a e e n e r e c G S n O
S U B C 2 I
r e v i r r D e v e i r n D i - L H
l r a e c v i t r i e r V D
o r e e v d r i i V D
R T t u p t u O H
r o s s e c o r P e s a b e m i T
c n y S _ V
y b d n a l y t S p p y r C u a D S i l i / x C r u A e A w o & P B S U
www.onsemi.com 37
s e i l p p u S r e w o P r o t i n o M ” 5 1 . 1 3 e r u g i F
c n y S _ H A T 8 h E F V C - S 0 N 0 O 6 M r e S . l l P o . t r M . n o S C . e C n . i L A
s e 0 1 2 8 0 c 5 5 6 6 6 i v 6 6 2 3 2 e 1 1 4 3 3 D P 3 3 3 P C C C C C C N M M M F N P
0 0 0 2 4 6 4 4 4 R R R U U U M M M
l c a n n y i g S S 5 0 5 x / 8 0 0 0 4 3 6 2 2 8 0 6 1 1 3 4 4 P P 4 4 C C C C C U N N M M
SMPSRM
Ultrafast Rectifier
Start-up Switch Rectifier
+
Bulk Storage Capacitor
AC Line
+
Load
PWM Control IC
MOSFET n-outputs
PWM Switcher
Prog. Prec. Ref
Figure 38. AC/DC Power Supply for CRT Displays
Table 1. Part #
Description
Key Parameters
Samples/Prod.
MC33262
PFC Control IC
Critical Conduction PFC Controller
Now/Now
MC33368
PFC Control IC
Critical Conduction PFC Controller + Internal Start-up
Now/Now
MC33260
PFC Control IC
Low System Cost, PFC with Synchronization Capability, Follower Boost Mode, or Normal Mode
Now/Now
MC33365
PWM Control IC
Fixed Frequency Controller + 700 V Start-up, 1 A Power Switch
Now/Now
MC33364
PWM Control IC
Variable Frequency Controller + 700 V Start-up Switch
Now/Now
MC44603A/604
PWM Control IC
GreenLine, Sync. Facility with Low Standby Mode
Now/Now
MC44605
PWM Control IC
GreenLine, Sync. Facility, Current-mode
Now/Now
MC44608
PWM Control IC
GreenLine, Fixed Frequency (40 kHz, 75 kHz and 100 kHz options), Controller + Internal Start-up, 8-pin
Now/Now
MSR860
Ultrasoft Rectifier
600 V, 8 A, trr = 55 ns, Ir max = 1 uA
Now/Now
MUR440
Ultrafast Rectifier
400 V, 4 A, trr = 50 ns, Ir max = 10 uA
Now/Now
MRA4006T3
Fast Recovery Rectifier
800 V, 1 A, Vf = 1.1 V @ 1.0 A
Now/Now
MR856
Fast Recovery Rectifier
600 V, 3 A, Vf = 1.25 V @ 3.0 A
Now/Now
NCP1200
PWM Current-Mode Controller
110 mA Source/Sink, O/P Protection, 40/60/110 kHz
Now/Now
NCP1205
Single-Ended PWM Controller
Quasi-resonant Operation, 250 mA Source/Sink, 8-36 V Operation
Now/Now
High Performance Current-Mode Controllers
500 kHz Freq., Totem Pole O/P, Cycle-by-Cycle Current Limiting, UV Lockout
Now/Now
UC3842/3/4/5
www.onsemi.com 38
SMPSRM
Ultrafast Rectifier
Start-up Switch Rectifier
+
Bulk Storage Capacitor
AC Line
+
Load
PWM Control IC
MOSFET n-outputs
PWM Switcher
Prog. Prec. Ref
Figure 39. AC/DC Power Supply for Storage, Imaging & Entertainment
Table 2. Part #
Description
Key Parameters
Samples/Prod.
MC33363A/B/65
PWM Control IC
Controller + 700 V Start-up & Power Switch, < 15 W
Now/Now
MC33364
PWM Control IC
Critical Conduction Mode, SMPS Controller
Now/Now
0.4% Tolerance, Prog. Output up to 36 V, Temperature Compensated
Now/Now
200 V, 6 A, trr = 55 ns, Ir max = 1 uA
Now/Now
600 V, 3 A, Vf = 1.25 V @ 3.0 A
Now/Now
TL431B MSRD620CT MR856
Program Precision Reference Ultrasoft Rectifier Fast Recovery Rectifier
NCP1200
PWM Current-Mode Controller
110 mA Source/Sink, O/P Protection, 40/60/110 kHz
Now/Now
NCP1205
Single-Ended PWM Controller
Quasi-resonant Operation, 250 mA Source/Sink, 8-36 V Operation
Now/Now
High Performance Current-Mode Controllers
500 kHz Freq., Totem Pole O/P, Cycle-by-Cycle Current Limiting, UV Lockout
Now/Now
UC3842/3/4/5
www.onsemi.com 39
SMPSRM Lo + V in
Lo
Voltage Regulation +
Co Control IC
+
Vout
-
Load
-
V in
+ Control IC
Co
Vout
-
Buck Regulator
Load
-
Synchronous Buck Regulator Figure 40. DC - DC Conversion
Table 3. Part #
Description
MC33263
Low Noise, Low Dropout Regulator IC
MC33269
Medium Dropout Regulator IC
MC33275/375 LP2950/51
Key Parameters
Low Dropout Regulator Low Dropout, Fixed Voltage IC
Samples/Prod.
150 mA; 8 Outputs 2.8 V - 5 V; SOT 23L 6 Lead Package
Now/Now
0.8 A; 3.3; 5, 12 V out; 1 V diff; 1% Tolerance
Now/Now
300 mA; 2.5, 3, 3.3, 5 V out
Now/Now
0.1 A; 3, 3.3, 5 V out; 0.38 V diff; 0.5% Tolerance
Now/Now
Iout = 150 mA, Available in 2.8 V, 3 V, 3.3 V, 5 V; SOT 23 - 5 Leads
Now/Now
MC78PC
CMOS LDO Linear Voltage Regulator
MC33470
Synchronous Buck Regulator IC
Digital Controlled; V cc = 7 V; Fast Response
Now/Now
NTMSD2P102LR2
P-Ch FET w/Schottky in SO-8
20 V, 2 A, 160 m W FET/1 A, Vf = 0.46 V Schottky
Now/Now
NTMSD3P102R2
P-Ch FET w/Schottky in SO-8
20 V, 3 A, 160 m W FET/1 A, Vf = 0.46 V Schottky
Now/Now
MMDFS6N303R2
N-Ch FET w/Schottky in SO-8
30 V, 6 A, 35 m W FET/3 A, Vf = 0.42 V Schottky
Now/Now
NTMSD3P303R2
P-Ch FET w/Schottky in SO-8
30 V, 3 A, 100 m W FET/3 A, Vf = 0.42 V Schottky
Now/Now
MBRM140T3
1A Schottky in POWERMITE Package
40 V, 1 A, Vf = 0.43 @ 1 A; Ir = 0.4 mA @ 40 V
Now/Now
MBRA130LT3
1A Schottky in SMA Package
40 V, 1 A, Vf = 0.395 @ 1 A; Ir = 1 mA @ 40 V
Now/Now
MBRS2040LT3
2A Schottky in SMB Package
40 V, 2 A, Vf = 0.43 @ 2 A; Ir = 0.8 mA @ 40 V
Now/Now
MMSF3300
Single N-Ch MOSFET in SO-8
30 V, 11.5 A (1), 12.5 mW @ 10 V
Now/Now
NTD4302
Single N-Ch MOSFET in DPAK
30 V, 18.3 A (1), 10 mW @ 10 V
Now/Now
NTTS2P03R2
Single P-Ch MOSFET in Micro8 Package
30 V, 2.7 A, 90 m W @ 10 V
Now/Now
MGSF3454X/V
Single N-Ch MOSFET in TSOP-6
30 V, 4.2 A, 65 m W @ 10 V
Now/Now
NTGS3441T1
Single P-Ch MOSFET in TSOP-6
20 V, 3.3 A, 100 m W @ 4.5 V
Now/Now
Prog. O/P Voltage 1.0, 1.3, 1.5, 1.8 V
Now/Now
NCP1500
Dual Mode PWM Linear Buck Converter
NCP1570
Low Voltage Synchronous Buck Converter
UV Lockout, 200 kHz Osc. Freq., 200 ns Response
Now/Now
NCP1571
Low Voltage Synchronous Buck Converter
UV Lockout, 200 kHz Osc. Freq., 200 ns Response
Now/Now
Dual Synchronous Buck Converter
150 kHz-600 kHz Prog. Freq., UV Lockout, 150 ns Transient Response
Now/Now
CS5422
(1) Continuous at T A = 25° C, Mounted on 1” square FR-4 or G10, V GS = 10 V t 10 seconds
www.onsemi.com 40
SMPSRM e 0 g 2 a k 2 c a O P T ) A ( 5 2 I o ) V ( M 5 R 3
e 0 0 0 0 8 7 e 0 0 0 0 0 0 e g 2 2 2 2 1 4 g 2 2 2 2 2 2 g l l a a 2 2 2 2 2 2 a C K l a i a i a 2 2 2 2 2 k A i k k 2 x x x c - - - - - - c - - - - - - c M P a O O O O O O a O O O O O O a S A A A P T T T T T T P T T T T T T P D ) ) ) A ( 5 5 0 5 0 0 A ( 0 0 0 6 6 6 A ( 2 2 3 2 3 3 o 2 2 2 1 1 1 o 3 3 3 3 3 I o I I ) ) ) V V V ( ( ( 0 0 0 0 M 0 0 0 0 0 M 5 5 5 5 5 5 M 0 0 0 0 0 0 0 R 3 4 4 4 4 4 R 6 1 2 2 2 2 R 4 4 3 4 4
V
V
R
R
V V L T T T T R T T T C T C C T T C . C C T . T . 3 5 S P W T C 0 C C 0 o 5 5 5 4 5 5 o 0 0 o 0 2 0 0 0 N 3 4 4 5 4 4 N 6 0 N 0 4 4 6 2 2 1 2 2 5 5 0 0 0 t t t 1 2 0 1 3 6 6 0 0 0 r 2 2 3 F 3 3 r 2 2 2 1 1 F r S 3 2 4 D 2 a R R R R R R a a 8 8 3 R R R R R R R R P B B B B B B P P 5 R B B B U U U B B 5 N N B M M M M M M M M 1 1 M M M M M M M
L T . C o 5 N 3 t r 5 a 2 P R B M A 4 1 V 3 . 3 +
e g l a a k i c x a A P
R
R
A 2 2 V 5 +
+
+
+
) V ( 0 M 0 R 1 R V . 0 o 0 N 1 3 t r R a B P M
A 8 . 0 V 2 1 -
A 5 . 0 V 5 -
A 6 V 2 1 +
e g l a a k i x c A a P ) A ( 3 I o
+
+
e g 2 a 9 k c O a P T . o N 1 3 t r 4 a L P T
Figure 41. . Typical 200 W ATX Forward Mode SMPS
X I R T A M
) A ( 1 I o ) V (
M 0 R 6 R
V
. o 0 6 N 1 t r R a B P M M C I W P
+
e g l l l a a i a i a k i c x x x a A A A P ) A ( 3 3 3 I o 0 0 ) 0 0 0 0 V ( 0 0 0 M 1 1 1
R X X X R 0 0 0
0 0 V 0 4 4 4
. L L L R o R R 6 8 N 4 0 0 0 t 4 4 r 4 a 5 5 5 N N P N 1 1 1
e g l l l l a a i a i a i a k i x x x x c A A A A a P ) A ( 1 4 6 1 I o ) 0 0 0 0 0 V ( 0 0 0 0 M 1 1 1 0 0 R X X X R 0 0 0 6 V 0 0 0 6 6 6
s c a n V i a 0 3 M 2
A 1 . 0 V 5
y e b g a t l d n o t a V S
4 1 - 4 6 6 6 6 6 1 1 1 1 1 e O 1 - - - - - g S a / O O O O 8 O O k 8 / S / S / S / P S / S / c - S 6 6 6 I 6 6 a O 4 D 1 1 1 1 1 1 P S / P I P I P I P I P I 8 I P D D D D D D P I D s e . i r A o e 0 3 8 3 3 N S 6 2 0 0 0 t 0 4 4 0 6 6 6 r X 4 9 9 4 4 4 4 a 4 3 5 3 4 4 4 P 8 3 C 4 L L C C C C U M T T M M M M
www.onsemi.com 41
E E 0 0 L 0 0 R 1 1 0 1 4 . R R 6 o U U 7 N M M R M t r , , , a E E L 0 7 P 0 8 8 R 1 4 6 5 3 R R 7 9 U U R 4 N M M M 1
S P M S e d o M d r a w r o F X T A W 0 0 2 l a c i p y T . 5 3 e r u g i F
SMPSRM
Application: 80 W Power Factor Controller 1
100 k R6 8
C5 MC33262
D2 R 92 to I E F T D L 1 138 Vac R I F
D4
+
ZERO CURRENT DETECTOR D3
+
1.6 V/ 1.4 V
2.5 V REFERENCE
TIMER
T
22 k R4
UVLO
10 DRIVE OUTPUT 10
20 k
CURRENT SENSE COMPARATOR
1.5 V OVERVOLTAGE COMPARATOR
MUR130 D5
7
500 V/8 A N-Ch MOSFET Q1
4 0.1 R7
10 pF
VO 230 V/ 0.35 A +
220 C3
1.0 M R2
+ 1.08 V
ref
MULTIPLIER
7.5 k R3
5
16 V
R
RS LATCH
0.01 C2
6.7 V
100 C4
+ 13 V/ 8.0 V
DELAY
2.2 M R5
+
36 V
1.2 V
1N4934 D6
ERROR AMP + 10 m A Vref
3
1
QUICKSTART 2 0.68 C1
6
Figure 42. 80 W Power Factor Controller
Features: Reduced part count, low-cost solution. ON Semiconductor Advantages: Complete semiconductor solution based around highly integrated MC33262. Devices:
Part Number
Description
MC33262 MUR130
Power Factor Controller Axial Lead Ultrafast Recovery Rectifier (300 V)
Transformer
Coilcraft N2881-A Primary: 62 turns of #22 AWG Secondary: 5 turns of #22 AWG Core: Coilcraft PT2510 Gap: 0.072” total for a primary inductance (Lp) of 320 mH
www.onsemi.com 42
11 k R1
SMPSRM
Application: Compact Power Factor Correction
Vcc FUSE
0.33 μF 1N5404
AC LINE 100 nF
10 μF/ 16 V
MAINS FILTER
L1
+
Vout
MUR460
+ 1 2
100 nF
3
8 0 6 2 3 3 C M
4
7 10 W
500 V/8 A N-Ch MOSFET
6 5
12 kW
1 MW
120 pF 45 kW
0.5 W /3 W
1 MW
Figure 43. Compact Power Factor Correction
Features : Low-cost system solution for boost mode follower. Meets IEC1000-3-2 standard. Critical conduction, voltage mode. Follower boost mode for system cost reduction - smaller inductor and MOSFET can be used. Inrush current detection. Protection against overcurrent, overvoltage and undervoltage. ON Semiconductor advantages: Very low component count. No Auxiliary winding required. High reliability. Complete semiconductor solution. Significant system cost reduction. Devices:
Part Number
Description
MC33260 MUR460 1N5404
Power Factor Controller Ultrafast Recovery Rectifier (600 V) General Purpose Rectifier (400 V)
www.onsemi.com 43
100 μF/ 450 V
SMPSRM
Application: Monitor Pulsed-Mode SMPS 90 Vac to 270 Vac 22 μH
1 nF/1 kV
RFI FILTER
MR856 1 nF/500 V
4.7 MW
1W
1 nF/500 V
150 μF 400 V
3.9 kW /6 W
47 μF
4.7 kW 1N4148 1N4934 MCR22-6
100 nF 1N4934
22 kW
MR856
+
47 μF 25 V 3.3 kW 1.2 kW
+
120 pF
2W SYNC
+
47 μF
Vin
D1 - D4 1N5404
90 V/0.1 A
9
8
10
7
10 pF
47 kW
45 V/ 1A
+
15 V/ 0.8 A
+
-10 V/ 0.3 A
+
8 V/ 1.5 A
1000 μF
MR856
1 μH
+
SMT31
2.2 nF
22 nF
4.7 μF 2.2 kW 11 + 8.2 kW 12 470 1N4148 kW 13 2.2 nF 14
56 kW
MR852
470 pF 6 P 5 0 6 4 4 C M
4.7 μF+ 10 V
5
4.7 μF+ 10 V
560 kW 4
1000 μF
Lp
150 kW 470 pF
MR852
Note 1
3
220 μF
10 W 15
2
1 kW
270 W 16
1 470 W
56 kW
1N4934
MBR360 0.1 W
10 kW
4700 μF 100 W
MOC8107 1.8 MW
96.8 kW
10 kW Vin
100 nF TL431 2.7 kW
1N4742A 12 V
2.7 kW
Note 1: 500 V/8 A N-Channel MOSFET 1 kW BC237B
100 nF VmP FROM mP 0: STAND-BY 1: NORMAL MODE
Figure 44. Monitor Pulsed-Mode SMPS
www.onsemi.com 44
SMPSRM Features: Off power consumption: 40 mA drawn from the 8 V output in Burst mode.
Vac (110 V) ³ Vac (240 V) ³
about 1 watt about 3 watts
Efficiency (pout = 85 watts) Around 77% @ Vac (110 V) Around 80% @ Vac (240 V) Maximum Power limitation. Over-temperature detection. Winding short circuit detection. ON Semiconductor Advantages: Designed around high performance current mode controller. Built-in latched disabling mode. Complete semiconductor solution. Devices:
Part Number
Description
MC44605P
High Safety Latched Mode GreenLine Controller For (Multi) Synchronized Applications Programmable Precision Reference Fast Recovery Rectifier (600 V) Fast Recovery Rectifier (200 V) Axial Lead Schottky Rectifier (60 V) NPN Bipolar Transistor General-Purpose Rectifier (400 V) Zener Regulator (12 V, 1 W)
TL431 MR856 MR852 MBR360 BC237B 1N5404 1N4742A Transformer
t
G6351-00 (SMT31M) from Thomson Orega Primary inductance = 207 mH Area = 190 nH/turns2 Primary turns = 33 Turns (90 V) = 31
www.onsemi.com 45
SMPSRM
Application: 70 W Wide Mains TV SMPS 95 Vac to 265 Vac F1 FUSE 1.6 A
C30 100 nF 250 Vac RFI FILTER
LF1
C19 1 nF/1 kV
R21 4.7 MW
D1-D4 1N4007
C1 220 mF
3.8 MW C4-C5 1 nF/1 kV
R7 68 kW /1 W
D13 1N4148 C16 100 μF
D15 1N4148 C9 100 nF
9 C8 560 pF 10
R3 22 kW
8 7
C10 1 μF 11 R18
12
5.6 kW R15 1 MW
C7 10 nF
13 14
6 P A 3 0 6 4 4 C M
5 4
15
C26 4.7 nF
D12 MR856 C20 47 μF
D5 MR854
C11 100 pF R22 C12 1 nF
16
L3 22 μH 115 V/0.45 A
D23 47 μF
15 V/1.5 A C21 1000 μF
1 kW 15 kW
11 V/0.5 A
180 kW
3 2
R8 1 kW
D8 MR854
Q1 600 V/4 A N-Ch MOSFET
1 R33 0.31 W
C14 220 pF
R13 10 kW
Figure 45. 70 W Wide Mains TV SMPS
www.onsemi.com 46
C22 1000 μF
OREGA TRANSFORMER G6191-00 THOMSON TV COMPONENTS
R9 150W
R5 2.2 kW R14 47 kW
C15 220 pF
D7 1N4937
L1 1 μH
R19 27 kW
R20 47W R4 3.9 kW
R16 68 kW /2 W
SMPSRM Features: 70 W output power from 95 to 265 Vac.
Efficiency @ 230 Vac = 86% @ 110 Vac = 84% Load regulation (115 Vac) = " 0.8 V. Cross regulation (115 Vac) = " 0.2 V. Frequency 20 kHz fully stable. ON Semiconductor Advantages: DIP16 or SO16 packaging options for controller. Meets IEC emi radiation standards. A narrow supply voltage design (80 W) is also available. Devices:
Part Number
Description
MC44603AP MR856 MR854 1N4007 1N4937
Enhanced Mixed Frequency Mode GreenLine PWM Controller Fast Recovery Rectifier (600 V) Fast Recovery Rectifier (400 V) General Purpose Rectifier (1000 V) General Purpose Rectifier (600 V)
Transformer
Thomson Orega SMT18
t
www.onsemi.com 47
SMPSRM
Application: Wide Mains 100 W TV SMPS with 1.3 W TV Stand-by
F1 C31 100 nF 47283900 R F6
C19 2N2F-Y
RFI FILTER
C3 1 nF
R16 4.7 MW /4 kV D1-D4 1N5404
C11 220 pF/500 V +
C4 1 nF D5 1N4007 R5 100 kW
C5 R1 220 mF 22 kW 400 V 5W
112 V/0.45 A 14
C6 47 nF 630 V
D6 MR856
6
16 V/1.5 A R7 47 kΩ C17 120 pF
D12 1N4934
Isense
7
8 5 7 P 8 0 6 4 4 C M
2 3
+
7 Vcc
6
4
5
R2 10 W
C8 100 nF
C7 22 mF 16 V
2
11
8 V/1 A
C14 + 1000 μF/35 V
3
10 C16 100 pF
8
D14 MR856
D10 MR852
9
R17 2.2 kW 5W
R4 3.9 k W
2
D9 MR852
C9 470 pF 630 V 600 V/6 A N-CH MOSFET
DZ1 MCR22-6
R19 18 kW
D13 1N4148
+
R12 1 kW
C15 1000 μF/16 V
C18 100 nF
R3 0.27 W ON
R21 47 W
OFF
R9 100 kW
OPT1
R11 47 kW
DZ3 10 V 1N4740A
C19 33 nF DZ2 TL431CLP
R10 10 kW
R8 2.4 kW
Figure 46. Wide Mains 100 W TV SMPS with Secondary Reconfiguration for 1.3 W TV Stand-by
www.onsemi.com 48
J3
3 1
1
D7 1N4148 1
2
C13 100 nF
C12 47 μF/250 V
12
1
+
D18 MR856
ON = Normal mode OFF = Pulsed mode
J4
SMPSRM Features: Off power consumption: 300mW drawn from the 8V output in pulsed mode. Pin = 1.3W independent of the mains. Efficiency: 83% Maximum power limitation. Over-temperature detection. Demagnetization detection. Protection against open loop. ON Semiconductor Advantages: Very low component count controller. Fail safe open feedback loop. Programmable pulsed-mode power transfer for efficient system stand-by mode. Stand-by losses independent of the mains value. Complete semiconductor solution. Devices:
Part Number
Description
MC44608P75 TL431 MR856 MR852 1N5404 1N4740A
GreenLine Very High Voltage PWM Controller Programmable Precision Reference Fast Recovery Rectifier (600 V) Fast Recovery Rectifier (200 V) General Purpose Rectifier (400 V) Zener Regulator (10 V, 1 W)
Transformer
SMT19 40346-29 (9 slots coil former) Primary inductance: 181 mH Nprimary: 40 turns N 112 V: 40 turns N 16 V: 6 turns N 8 V: 3 turns
t
www.onsemi.com 49
SMPSRM
Application: Low-Cost Offline IGBT Battery Charger +
130 to 350 V DC R1
D1
150
1N4148
C3 220 mF/ 10 V
+
C2 220 mF/ 10 V
D3 R3 220 k
C10 1 nF
R13 100 k
R1
150
1N4148
M1 MMG05N60D
R11 113 k
120 k
C3 10 mF/ 350 V
R5
IC1 MOC8103
1k MC14093 R5 1.2 k
8
7
6
5
+ 1
D2 12 V
R9
C4 47 nF
470 C5 1 nF
Q1 MBT3946DW
C9 1 nF
2
R2 3.9
R9 100
Q5 R10
0V
Figure 47. Low-Cost Offline IGBT Battery Charger
Features: Universal ac input. 3 Watt capability for charging portable equipment. Light weight. Space saving surface mount design. ON Semiconductor Advantages: Special-process IGBT (Normal IGBTs will not function properly in this application). Off the shelf components. SPICE model available for MC33341. Devices:
Part Number
Description
MMG05N60D MC33341 MBT3946DW MBRS240LT3 MC14093 1N4937
Insulated Gate Bipolar Transistor in SOT-223 Package Power Supply Battery Charger Regulator Control Circuit Dual General Purpose (Bipolar) Transistors Surface Mount Schottky Power Rectifier Quad 2-Input “NAND” Schmitt Trigger General-Purpose Rectifier (600 V)
www.onsemi.com 50
3
+
C8 1 mF
MC33341
C7 10 mF
-
MBRS240LT3 D5 R2
D4 1N4937
+
8 V at 400 mA
+
4
D4 12 V R12 20 k
SMPSRM
Application: 110 W Output Flyback SMPS 180 VAC TO 280 VAC
RFI FILTER
C3 1 nF / 1 KV
R1 1W/5W
R3 4.7 kW
C4-C7 1 nF / 1000 V
C32 C1 100 mF
D1-D4 1N4007 C2 220 mF R2 68 kW / 2 W 9 C9 C10
820 pF 1 mF
R15 10 kW C11 1 nF R16 10 kW
R18 27 kW
10
7
11
6
12 13
P 3 0 6 4 4 C M
L1 1 mH
28 V / 1 A D9 MR852
3
15
2
16
1
R19 10 kW
R6 180 W
C26
220 pF
D10 MR852
R26 1 kW
C25 1000 mF
C23
C24 0.1 mF
220 pF
R14 2 X 0.56 W //
8V/1A D11 MR852 C21 1000 mF
R17 10 kW
R24 270 W
R21 C19 10 kW 100 nF
R25 1 kW
C28 0.1 mF
15 V / 1 A
R10 10 W
C13 100 nF
C27 1000 mF
LP C14 4.7 nF
Note 1 14
C31 0.1 mF
220 pF
R5 1.2 kW
R8 15 kW
4
C29
D6 1N4148
C15 1 nF R7 180 kW
D7 MR856
C30 100 mF
Laux
R9 C16 100 pF 1 kW
5
D8 MR856
C17 47 nF
R4 27 kW 8
120 V / 0.5 A
R20 22 kW 5W
D5 1N4934
220 pF
C12 6.8 nF
TL431 Note 1: 600 V/ 6 A N-Channel MOSFET
Figure 48. 110 W Output Flyback SMPS
www.onsemi.com 51
C22 0.1 mF
R23 117.5 kW
D14 1N4733
C20 33 nF R22 2.5 kW
SMPSRM Features: Off-line operation from 180 V to 280 Vac mains. Fixed frquency and stand-by mode. Automatically changes operating mode based on load requirements. Precise limiting of maximum power in fixed frequency mode. ON Semiconductor Advantages: Built-in protection circuitry for current limitation, overvoltage detection, foldback, demagnetization and softstart. Reduced frequency in stand-by mode. Devices:
Part Number
Description
MC44603P MR856 MR852 TL431 1N4733A 1N4007
Enhanced Mixed Frequency Mode GreenLine Fast Recovery Rectifier (600 V) Fast Recovery Rectifier (200 V) Programmable Precision Reference Zener Voltage Regulator Diode (5.1 V) General Purpose Rectifier (1000 V)
t
www.onsemi.com 52
PWM Controller
SMPSRM
Application: Efficient Safety Circuit for Electronic Ballast C13 100 nF
C14 100 nF AGND
250 V
250 V
C12 22 nF
R18 PTC
C11 4.7 nF 1200 V PTUBE = 55 W
T1A FT063
L1 1.6 mH
Q3 MJE18004D2
Q2 MJE18004D2 R13 2.2 R
R14 2.2 R
R11 4.7 R
C9 2.2 nF
C8 2.2 nF
R12 4.7 R
DIAC C6 10 nF
C7 10 nF
NOTES: * All resistors are ± 5%, 0.25 W unless otherwise noted * All capacitors are Polycarbonate, 63 V, ± 10%, unless otherwise noted
D4 R10 10 R
T1B D3 1N4007
T1C C5 0.22 mF
R9 330 k
C4 47 mF + 450 V
R7 1.8 M
P1 20 k C15 100 nF
D2 MUR180E
Q1 500 V/4 A N-Ch R6 1.0 R MOSFET 3
1 2
T2
1N5407
D8
D9 C16 47 nF
R5 1.0 R
AGND
R4 22 k 5 + D1 MUR120
1N5407
C2 330 mF 25 V
8 R3 100 k/1.0 W R2 1.2 M
7
D7 1N5407
4
2 2 6 1 2 4 6 3 U C M 1
3
630 V 1N5407 D6 FILTER
C3 1.0 mF
C17 47 nF 630 V
C1 10 nF FUSE LINE 220 V
R1 12 k
Figure 49. Efficient Safety Circuit for Electronic Ballast
www.onsemi.com 53
SMPSRM Features: Easy to implement circuit to avoid thermal runaway when fluorescent lamp does not strike. ON Semiconductor Advantages: Power devices do not have to be oversized - lower cost solution. Includes power factor correction. Devices:
Part Number
Description
MC34262 MUR120 MJE18004D2 1N4007 1N5240B 1N5407
Power Factor Controller Ultrafast Rectifier (200 V) High Voltage Planar Bipolar Power Transistor (100 V) General Purpose Diode (1000 V) Zener Voltage Regulator Diode (10 V) Rectifier (3 A, 800 V)
*Other Lamp Ballast Options:
825 V 100 V 1200 V
1, 2 Lamps
3, 4 Lamps
BUL642D2 MJD18002D2 MJD18202D2
BUL642D2 MJB18004D2 MJB18204D2 MJE18204D2
ON Semiconductor's H2BIP process integrates a diode and bipolar transistor for a single package solution.
www.onsemi.com 54
SMPSRM
Application: AC-DC Battery Charger - Constant Current with Voltage Limit T0.2x
J1
D1
F1
1
250R 1N4140
+
2 LINE
C1
10 mF/350 V 10 V
R1
1N4140
220
D3 4 kW D4
C2 20 mF
BZX84/18V
1N4140 R3
8
7
1
Line
VCC
ICD
U1
5
Vref C3 100 nF
C3 FL
4
22 k 1N4140 6
MC33364 GND
2
4
R2
+ D2
5
D5
R8 100 T1 6
R5
7
3
R4
D8 C5 + 4 kW D9 1 mF BZX84/5 V 100 mF + D7
1 2
R14 22 k U2
MURS160T3
8
7 6 5 I I O C S S D C C V V MC33341 I A P D S T M N C C C G
Q1 600 V/1 A N-Ch MOSFET
330
J2
C5
MURS320T3
47 k R6 47 k C4 1 nF 2 D6
R4
5V
R7 2.7
R13 12 k
1 2 3 4 C7
3 5
1SO1
2
MOC0102 4
R10 100 R
1
33 nF R11 0.25
Figure 50. AC-DC Battery Charger - Constant Current with Voltage Limit
Features: Universal ac input. 9.5 Watt capability for charging portable equipment. Light weight. Space saving surface mount design. ON Semiconductor Advantages: Off the shelf components SPICE model available for MC33341 Devices:
Part Number
Description
MC33341 MC33364 MURS160T3 MURS320T3 BZX84C5V1LT1 BZX84/18V Transformer
Power Supply Battery Charger Regulator Control Circuit Critical Conduction SMPS Controller Surface Mount Ultrafast Rectifier (600 V) Surface Mount Ultrafast Rectifier (200 V) Zener Voltage Regulator Diode (5.1 V) Zener Voltage Regulator Diode (MMSZ18T1) For details consult AN1600
www.onsemi.com 55
R12 10 k
SMPSRM
Literature Available from ON Semiconductor Application Notes These older Application Notes may contain part numbers that are no longer available, but the applications information may still be helpful in designing an SMPS. They are available through the ON Semiconductor website at www.onsemi.com.
AN873 - Understanding Power Transistor Dynamic Behavior: dv/dt Effects on Switching RBSOA AN875 - Power Transistor Safe Operating Area: Special Consideration for Switching Power Supplies AN913 - Designing with TMOS Power MOSFETs AN915 - Characterizing Collector-to-Emitter and Drain-to-Source Diodes for Switchmode Applications AN918 - Paralleling Power MOSFETs in Switching Applications AN920 - Theory and Applications of the MC34063 and m A78S40 Switching Regulator Control Circuits AN929 - Insuring Reliable Performance from Power MOSFETs AN952 - Ultrafast Recovery Rectifiers Extend Power Transistor SOA AN1040 - Mounting Considerations for Power Semiconductors AN1043 - SPICE Model for TMOS Power MOSFETs AN1080 - External-Sync Power Supply with Universal Input Voltage Range for Monitors AN1083 - Basic Thermal Management of Power Semiconductors AN1090 - Understanding and Predicting Power MOSFET Switching Behavior AN1320 - 300 Watt, 100 kHz Converter Utilizes Economical Bipolar Planar Power Transistors AN1327 - Very Wide Input Voltage Range, Off-line Flyback Switching Power Supply AN1520 - HDTMOS Power MOSFETs Excel in Synchronous Rectifier Applications AN1541 - Introduction to Insulated Gate Bipolar Transistor AN1542 - Active Inrush Current Limiting Using MOSFETs AN1543 - Electronic Lamp Ballast Design AN1547 - A DC to DC Converter for Notebook Computers Using HDTMOS and Synchronous Rectification AN1570 - Basic Semiconductor Thermal Measurement AN1576 - Reduce Compact Fluorescent Cost with Motorola's (ON Semiconductor) IGBTs for Lighting AN1577 - Motorola's (ON Semiconductor) D2 Series Transistors for Fluorescent Converters AN1593 - Low Cost 1.0 A Current Source for Battery Chargers AN1594 - Critical Conduction Mode, Flyback Switching Power Supply Using the MC33364 AN1600 - AC-DC Battery Charger - Constant Current with Voltage Limit
www.onsemi.com 56
SMPSRM
Literature Available from ON Semiconductor (continued) AN1601 - Efficient Safety Circuit for Electronic Ballast AN1628 - Understanding Power Transistors Breakdown Parameters AN1631 - Using PSPICE to Analyze Performance of Power MOSFETs in Step-Down, Switching Regulators Employing Synchronous Rectification AN1669 - MC44603 in a 110 W Output SMPS Application AN1679 - How to Deal with Leakage Elements in Flyback Converters AN1680 - Design Considerations for Clamping Networks for Very High Voltage Monolithic Off-Line PWM Controllers AN1681 - How to Keep a Flyback Switch Mode Supply stable with a Critical-Mode Controller Brochures and Selector Guides The following literature is available for downloading from the ON Semiconductor website at www.onsemi.com . Master Components Selector Guide
SG388/D
Power Conversion Solutions Selector Guide
SGD510/D
Power Supply & Power Adapter Solutions
BRD8063/D
Device Models Device models for SMPS circuits (MC33363 and MC33365), power transistors, rectifiers and other discrete products are available through ON Semiconductor's website or by contacting your local sales office.
www.onsemi.com 57
SMPSRM
Reference Books Relating to Switching Power Supply Design Baliga, B. Jayant, Power Semiconductor Devices, PWS Publishing Co., Boston, 1996. 624 pages. Brown, Marty, Practical Switching Power Supply Design, Academic Press, Harcourt Brace Jovanovich, 1990. 240 pages. Brown, Marty Power Supply Cookbook, EDN Series for Design Engineers, ON Semiconductor Series in Solid State Electronics, Butterworth-Heinmann, MA, 1994. 238 pages Chrysiss, G. C., High Frequency Switching Power Supplies: Theory and Design, Second Edition, McGraw-Hill, 1989. 287 pages Gottlieb, Irving M., Power Supplies, Switching Regulators, Inverters, and Converters, 2nd Edition, TAB Books, 1994. 479 pages. Kassakian, John G., Martin F. Schlect, and George C. Verghese, Principles of Power Electronics, Addison-Wesley, 1991. 738 pages. Lee, Yim-Shu, Computer-Aided Analysis and Design of Switch-Mode Power Supplies, Marcel Dekker, Inc., NY, 1993 Lenk, John D., Simplified Design of Switching Power Supplies, EDN Series for Design Engineers, Butterworth-Heinmann, MA, 1994. 221 pages. McLyman, C. W. T., Designing Magnetic Components for High Frequency DC-DC Converters, KG Magnetics, San Marino, CA, 1993. 433 pages, 146 figures, 32 tables Mitchell, Daniel, Small-Signal MathCAD Design Aids, e/j Bloom Associates, 115 Duran Drive, San Rafael, Ca 94903-2317, 415-492-8443, 1992. Computer disk included. Mohan, Ned, Tore M. Undeland, William P. Robbins, Power Electronics: Converter, Applications and Design, 2nd Edition, Wiley, 1995. 802 pages Paice, Derek A., Power Electronic Converter Harmonics, Multipulse Methods for Clean Power, IEEE Press, 1995. 224 pages. Whittington, H. W., Switched Mode Power Supplies: Design and Construction, 2nd Edition, Wiley, 1996 224 pages. Basso, Christophe, Switch-Mode Power Supply SPICE Cookbook, McGraw-Hill, 2001. CD-ROM included. 255 pages.
www.onsemi.com 58
SMPSRM
Web Locations for Switching-Mode Power Supply Information Ardem Associates (Dr. R. David Middlebrook) http://www.ardem.com/ Applied Power Electronics Conference (APEC) The power electronics conference for the practical aspects of power supplies. http://www.apec-conf.org/ Dr. Vincent G. Bello's Home Page SPICE simulation for switching-mode power supplies. http://www.SpiceSim.com/ e/j BLOOM Associates (Ed Bloom) Educational Materials & Services for Power Electronics. http://www.ejbloom.com/ The Darnell Group (Jeff Shepard) Contains an excellent list of power electronics websites, an extensive list of manufacturer's contact information and more. http://www.darnell.com/ Switching-Mode Power Supply Design by Jerrold Foutz An excellent location for switching mode power supply information and links to other sources. http://www.smpstech.com/ Institute of Electrical and Electronics Engineers (IEEE) http://www.ieee.org/ IEEE Power Electronics Society http://www.pels.org/pels.html Power Control and Intelligent Motion (PCIM) Articles from present and past issues. http://www.pcim.com/ Power Corner Frank Greenhalgh's Power Corner in EDTN http://fgl.com/power1.htm Power Designers http://www.powerdesigners.com/ Power Quality Assurance Magazine Articles from present and past issues. http://powerquality.com/ Power Sources Manufacturers Association A trade organization for the power sources industry. http://www.psma.com/ Quantum Power Labs An excellent hypertext-linked glossary of power electronics terms. http://www.quantumpower.com/ Ridley Engineering, Inc. Dr. Ray Ridley http://www.ridleyengineering.com/
www.onsemi.com 59
SMPSRM
Web Locations for Switching-Mode Power Supply Information (continued) Springtime Enterprises - Rudy Severns Rudy Severns has over 40 years of experience in switching-mode power supply design and static power conversion for design engineers. http://www.rudyseverns.com/ TESLAco Dr. Slobodan Cuk is both chairman of TESLAco and head of the Caltech Power Electronics Group. http://www.teslaco.com/ Venable Industries http://www.venableind.com/
www.onsemi.com 60
Sales and Design Assistance from ON Semiconductor NORTH AMERICA SALES OFFICES & REP FIRMS Alabama
www.onsemi.com
Huntsville
e-Components
(256) 533-2444
Huntsville
ON Sales Ofce
(256) 774-1004
Arizona
Phoenix
Centaur Corporation
(480) 839-2320
Caliornia
Irvine
Centaur Corporation
(949) 261-2123
Los Angeles
Centaur Corporation
(805) 579-0519
San Diego
Centaur Corporation
(858) 278-4950
CANADA
All Regions
Astec
(905) 6 07-1444
Colorado
Englewood
ON Sales Ofce
Karl Hill: (720) 895-1367 Mike Barone: (303) 670-8000
Connecticut
Monroe
Impact Technical Sales
(203) 268-7229
Florida
Boca Raton
e-Components
(561) 289-1379
Lake Mary
e-Components
(407) 804-0042
Plantation
ON Sales Ofce
(954) 325-1223
Palm Harbor
e-Components
(727) 937-3016
Melbourne
e-Components
(321) 676-5659
Georgia
Duluth
e-Components
(678) 380-5080
Illinois
Homan Estates
Stan Clothier
(847) 781-4010
Indiana
Fishers Kokomo
Bear VAI ON Sales Ofce
(317) 570-0707 (765) 865-2091 (319) 393-1576
ON Semiconductor Distribution Partners Arro w Elect ronics
www.a rrow.co m
(80 0) 777-2776
Iowa
Cedar Rapids
Stan Clothier
Avnet
www.e m.avne t.com
(80 0) 332- 8638
Kansas
Overland Park
Stan Clothier
(913) 894-1675
Maryland
Columbia
Third Wave Solutions
(410) 290-5990
Massachusetts
Burlington
Impact Technical Sales
(781) 238-8888
Lowell
ON Sales Ofce
(978) 452-6075
Michigan
Livonia
ON Sales Ofce
(734) 953-6848
Minnesota
Eden Prairie
Stan Clothier
(952) 944-3456
Missouri
St. Charles
Stan Clothier
(636) 916-3777
New Jersey
Denville
CJR Associates
(973) 625-2796
New York
Long Island
CJR Associates
(973) 625-2796
New York City
CJR Associates
(973) 625-2796
Rochester
TriTech
(716) 385-6500 (607) 722-3580
Chip Supply
www.chipsupply.com
(407) 298-7100
Digi-Key
www.digikey.com
(800) 338-1003
EBV Elektronik
w ww.ebv.com/en/locations.html
(49) 8121 774-0
Farnell InOne
www.arnellinone.com
+44 8701 200 200
Future Electronics Europe
www.utureelectronics.com/contact
ino-eur-uture@utureelectronics.com
Future & FAI Electronics
www.utureelectronics.com/contact
1-800-FUTURE1 (388-8731)
Marubun/Arrow
www.marubunarrow.com
(852) 2375-1126
Minco
www.mincotech.com
(512) 834-2022
Mouser Electronics
www.mouser.com
(800) 346-6873
Newark Electronics
www.newark.com
(800) 4-NEWARK
NU Horizons
www.nuhorizons.com
(888) 747-6846
NuVision
www.nuvision-tech.com
Binghamton
TriTech
Ohio
Brecksville
Bear VAI
(937) 748-1341
(886) 2 8228-0688
Oregon
Portland
SiFore Technical Sales
(503) 977-6267
Rochester Electronics
www.rocelec.com
(978) 462-9332
Pennsylvania
Hatboro
Delta Technical Sales
(215) 957-0600
Semi Dice Inc
www.semidice.com
1 (800) 345-6633
Texas
Austin
West As sociate s
(512) 343 -1199
Silica (An Avnet Company)
www.silica.com
+33 1 6447 2929
Richardson
West Associates
(972) 680-2800
WPI
www.wpi-group.com
(886) 2 2788-5200
Washington
Bellevue
SiFore Technical Sales
(425) 990-4701
+49 6103 3040
Wisconsin
Statewide
Stan Clothier
(847) 781-4010
Spoerle (An Arrow Company) www.spoerle.de/de/about/ofces
INTERNATIONAL BRAZIL GREATER CHINA
S an Paulo Beijing Chengdu Hong Kong Shenzhen Shanghai Taipei, Taiwan
INTERNATIONAL ( 55 -11) 3 842-8 911 86-10-8518-2323 86-28-8678-4078 852-2689-0088 86-755-8209-1128 86-21-5131-7168 886 -2-2377-9911
FRANCE GERMANY INDIA ISRAEL ITALY JAPAN KOREA
Paris Munich Bangalore Herzelia Milan Tokyo Seoul
INTERNATIONAL 33 (0)1 39-26-41-00 49 (0) 89-93-0808-0 91- 80 -2532-50 84 972 (0) 9-9609-111 39 02-530-0951 81-3-5773-3 850 82-2-2190-3500
MALAYSIA MEXICO SINGAPORE SLOVAKIA UNITED KINGDOM
Penang Guadalajara Singapore Piestany Slough
60-4-226-9368 (5233) 3123-9199 65-6391-1260 421 33 790 2450 44 (0) 1753 70 1676 May-07
ON Semiconductor and the ON logo are registered trademarks o Semiconductor Components Industries, LLC (SCILLC ). SCILLC reserves the right to make changes without urther notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability o its products or any particular purpose, nor does SCILLC assume any liability arising out o the application or use o any product or circuit, and specifcally disclaims any and all liability, including without limitation special, consequential or incidental damages. “Typical” parameters which may be provided in SCILLC data sheets and/or specifcations can and do vary in dierent applications and actual perormance may vary over time. All operating parameters, including “Typicals” must be validated or each customer application by customer’s technical experts. SCILLC does not convey any license under its patent rights nor the rights o others. SCILLC products are not designed, intended, or authorized or use as components in systems intended or surgical implant into the body, or other applications inte nded to support or sustain lie, or or any other applicati on in which the ailure o the SCILLC produc t could create a situation where personal injur y or death may occur. S hould Buyer purchas e or use SCILLC product s or any such unintended or unauthorized application, Buyer shall indemniy and hold SCIL LC and its o fcers, employees, subsidiaries, afliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney ees arising out o, directly or indirectly, any claim o personal injury or death associated with such unintended or unauthorized use, even i such claim alleges that SCILLC was negligent regarding the design or manuacture o the part. SCILLC is an Equal Opportunity/Afrmative Action Employer. This literature is subject to all applicable copyright laws and is not or resale in any manner.
PUBLICATION ORDERING INFORMATION LITERATURE FULFILLMENT: Literature Distribution Center for ON Semiconductor P.O. Box 5163, Denver, Colo rado 80217 USA Phone: 303-675-2175 or 800-344-3860 Toll Free USA/Canada Fax: 303-675-2176 or 800-344-3867 Toll Free USA/Canada Email:
[email protected] PDF
N. American Technical Support : 800-282-9855 Toll Free USA/Canada. Europe, Middle East and Africa Technical Support : Phone: 421 33 790 2910 Japan Customer Focus Center Phone: 81-3-5773-3850
ON Semiconductor Website : www.onsemi.com Order Literature : http://www.onsemi.com/orderlit
For additional information, please contact your l ocal Sales Representative
