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AN1715 - APPLICATION NOTE VIPower: SMPS Solutions for Power Line Modem Application with VIPerX2A F. Cacciotto - F. Gennaro - M. Sciortino
1. ABSTRACT This application note investigates about possible power supply solutions based on VIPerX2A family, realized in order to power a Power Line Modem System (PLMS). As a starting point, the power supplies have been designed and developed according to the specifications for a complete PLMS based on ST7538 (by STMicroelectronics ), ), but other diffused PLMS can be suitably supplied.
2. INTRODUCTION The growth of the automation system in home appliance has brought the development of systems able to exchange information using the electrical network as a communication medium. As a result, there is no need to install extra control cable and all the system components can be connected to the network by plugging them in a wall socket. These virtual networks also improve the flexibility and the expansibility of the system, since new devices can be instantly connected to the system by means of a wall socket. New dedicated modem integrated circuits have been developed in order to make these applications feasible. A typical PLMS is shown in figure 1. The Power Line Modem (PLM) is a half duplex asynchronous FSK modem with a carrier frequency complying with Europe’s CENELEC EN50065 standard, which specifies the use of carrier frequencies from 125kHz to 140kHz for home automation and US FCC regulations, which specifies the use of carrier frequencies lower than 450kHz. The Power Line Interface (PLI) connects the PLM to the power lines. It consists in a line driver, which amplifies the Analog Transmit Output signal (ATO) from the PLM and a line interface, which adapts the line driver to the power line and insulates the PLMS from the electrical network. Some PLMs directly integrate the line interface on the chip. The PLI has the following functions: - TX Mode: amplifies and filters the transmit signal from the ATO; - RX Mode: provides received signal from power lines to the Receive Analog Input (RAI). The PLM is connected to a microcontroller or to a Personal Computer (through the RS232 driver interface), in order to build a home LAN, where each device is able to use any information required whether it is local (washing machine) or remote (remote control system). In the previous typical application, the power supply has to be able to provide a single output.
3. VIPerX2A DESCRIPTION The VIPerX2A family is a range of smart power devices with current mode PWM controller, start-up circuit and protections integrated in a monolithic chip using VIPower M0 Technology.
May 2003
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AN1715 - APPLICATION NOTE Figure 1: Power Line Modem: System configuration
Mains 50/60 Hz
PLMS
PLMS
Switch Mode Power Supply +10V
Washing machine
+5V +5V
DV CC
AV CC
Remote Control System
+5V LDO Regulator
µC
Power Line Modem
Power Driver
Analog Front End
Line interface
Power Line Modem System (PLMS)
The VIPerX2A family includes: - VIPer12A, with a 0.4A peak drain current limitation and 730V breakdown voltage; - VIPer22A, with a 0.7A peak drain current limitation and 730V breakdown voltage. The switching frequency is internally fixed at 60kHz by the integrated oscillator of the VIPerX2A. The internal control circuit offers the following benefits: - Large input voltage range on the VDD pin accommodates changes in auxiliary supply voltage; - Automatic burst mode in low load condition; - Overtemperature, overcurrent and overvoltage protection with auto-restart. The internal current mode structure is shown in figure 2. The feedback pin (FB pin) is sensitive to current and controls the operation of the device. The Power MOSFET delivers a sense current IS which is proportional to the drain current ID. R2 receives this current and the current coming from the FB pin. The voltage across R2 is then compared to a fixed reference voltage of 0.23 V.
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AN1715 - APPLICATION NOTE Figure 2: VIPerX2A current internal structure
DRAIN 60kHz Oscillator
ID
S
PWM Q LATCH
+VDD
R
IS
Secondary feedback 0.23V IFB
1k Ω
FB
R1 230Ω
R2
SOURCE
The mosfet is switched off when the following condition is reached:
R 2 ⋅ (I S
+ I FB ) = 0.23V
(1)
Using the current sense ratio of the mosfet, GID and considering (1), ID is given by: ID
= G ID ⋅ I S = G ID ⋅ 0.23V − I FB R 2
(2)
The FB pin is commonly driven by the emitter of an optocoupler but a discrete BJT or a zener diode can also be used, behaving as a current source. This current is filtered by a small capacitor C to guarantee the feedback stability. It is necessary to keep this capacitor very close to the FB pin, to avoid high frequency instability on the compensation loop. For low drain currents, (2) applies as long as IFB
=
(V
IN DC(min)
⋅ t ON(min) )2
2⋅L
⋅ f SW
(3)
This feature is especially important when the converter is lightly loaded, in order to have very low input power consumption.
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AN1715 - APPLICATION NOTE 4. POWER SUPPLY DESCRIPTION AND DESIGN The SMPS specifications are listed in table 1.
Table 1: SMPS specifications. 185V AC ÷ 265VAC 10V ± 25% 23mA in RX mode 480mA in TX mode
Input Voltage VOUT IOUT(min) IOUT(max)
Due to the low power related to the RX mode, as low as possible switching frequency can be chosen, in order to have higher order harmonics in the carrier frequency band. The only way to reduce the switching frequency is to optimise the burst mode operation.
4.1. Clamp Design The drain voltage needs to be clamped in order to prevent voltage spikes, due to leakage inductance, from exceeding the breakdown voltage of the device (730V minimum). The most used solution is the RCD clamp, as shown in figure 3a. This is a very simple and cheap solution, but it impacts on the efficiency even at no load condition. If the standby efficiency is important, a zener clamp is recommended, as shown in figure 3b. However such a solution gives higher power dissipation at full load, even if the clamp voltage is exactly defined.
Figure 3: Clamp circuit topology: (a) RCD clamp and (b) Zener clamp
RCD CLAMP
ZENER CLAMP
.
R
.
C
Dz +
+
D
D
(a)
(b)
The capacitor value is calculated in order to charge it with the energy from the leakage inductance and must ensure that the maximum VSPIKE is never exceeded, thus from energy balance consideration, the minimum capacitance value is:
C≥
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L LK ⋅ I 2DLIM (VSPIKE
(4)
+ VR ) − V 2
2 R
AN1715 - APPLICATION NOTE In order to have a proper operation of the clamp, the minimum value of resistance is: 1 R≥ V f SW ⋅ C ⋅ ln1 + SPIKE VR
(5)
Its power rating will be:
PR
=
VR2 R
1
+ ⋅ L LK ⋅ I 2DLIM ⋅ f SW
(6)
2
For a zener clamp, the zener voltage should be: VZ
= VR + VSPIKE
(7)
with a power capability equal to: VZ 1 PZ = ⋅ ⋅ L LK ⋅ I 2DLIM ⋅ f SW 2 VZ − VR
(8)
5. APPLICATION DESCRIPTION In this chapter, two solutions are presented in order to power a typical PLMS in both isolated and non isolated applications. The first configuration is typical in home automation systems and the last is suitable for many industrial applications. The regulation is obtained by means of a zener diode in either solution, considering the high output voltage tolerance for this application. The transformer has been designed with lower primary inductance compared to a typical 5W application. This enables the device to work in burst mode during RX condition, reducing the average switching frequency.
5.1. Isolated Solution The first proposed solution regards the isolated Flyback topology with a single input rectifier diode and an input C-L-C filter. Such a filter provides both DC voltage stabilization and EMI filtering. In the considered application, the transformer has a secondary winding with galvanic insulation and an auxiliary winding to supply the VIPer. In table 2 the transformer specifications are listed and the converter schematic is shown in figure 4.
Table 2: Isolated transformer specifications Core geometry Core material BSAT Air Gap Primary Inductance Leakage Inductance Primary Winding Auxiliary Winding Output Winding
E13/7/14 N27 or equivalent 380mT 0.24mm 1.8mH 54µH 166 turns 52 turns 22 turns
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AN1715 - APPLICATION NOTE The converter has been tested in several load conditions and in the whole input voltage range, i.e. from 185VAC to 265VAC. Load and line regulation results are shown in figure 5 and figure 6 respectively. The efficiency measurement has been done using a DC power source and an amperometer, in order to have a higher accuracy than in AC measurements: the results are shown in figure 7. In all the considered operating conditions, the converter meets the specifications given in table 1.
Figure 4: Isolated Flyback converter
R-fuse
D1
L1
1N4007
470uH
AC IN 10
TF1
1.2k
C6 470uF
STTH106 D2
+10V
L2 22uH
D4
C5 100pF
R2 150k
R1
BYW100-200
+
C7 47uF
+
D3 BAS21 GND OUT +
C1 4.7uF
+
C2 4.7uF
FB
+
C3 10uF
R4 220
DRAIN
VDD
CONTROL
OPT PC817 SOURCE
VIPer12A
C4 47nF
C8 2.2nF-2kV/Y2
AC IN
Figure 5: Load Regulation
9.8 185V 220V 265V
9.6
9.4 ) V (
T U O
V9.2
9.0
8.8 0
100
200
300
IOUT (mA)
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400
500
DZ1 8.2V
AN1715 - APPLICATION NOTE Figure 6: Line Regulation
10,0 Rx Mode Full load
9,5 ) V (
T U O
V
9,0
8,5 180
200
220
240
260
280
VINAC(V)
Figure 7: Efficiency Vs. output current
100 185V 230V 265V
90
80 ) % ( y 70 c n e i c i f f E 60
50
40 0
100
200
300
400
500
IOUT (mA)
In figures. 8, 9 and 10 typical waveforms in RX mode and full load are shown: it is important to point out that, in RX Mode, the converter works in burst mode, limiting the maximum switching frequency to 30kHz. The startup transient is shown in figure 11. The maximum drain voltage has been measured under worstcase operations, i.e. start-up at VIN=265VAC and full load. The maximum measured value is 594V, as shown in figure 11(b) and the output voltage ripple at full load and VIN=230VAC is shown in figure 12.
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AN1715 - APPLICATION NOTE Figure 8: Typical waveforms at 185VAC: (a) RX mode and (b) Full load
Ch1 ∆Freq – 29.41kHz
Ch1 Freq – 58.40kHz
Ch1 Freq – 14.65kHz Ch1 Max – 376V Ch2 Max – 140mA
Ch1 Max –444V Ch2 Max – 304mA
(b)
(a)
Figure 9: Typical waveforms at 230VAC: (a) RX mode and (b) Full load
Ch1 ∆Freq – 29.41kHz
Ch1 Freq – 58.19kHz Ch1 Max –504V Ch2 Max – 296mA
Ch1 Freq – 14.67kHz Ch1 Max – 444V Ch2 Max – 156mA
(a)
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(b)
AN1715 - APPLICATION NOTE Figure 10: Typical waveforms at 265VAC: (a) RX mode and (b) Full load
Ch1 ∆Freq – 29.41kHz Ch1 Freq – 14.65kHz
Ch1 Freq – 58.22kHz Ch1 Max –560V Ch2 Max – 296mA
Ch1 Max – 500V Ch2 Max – 164mA
(b)
(a)
Figure 11: (a) Start up time at 230VAC and (b) VDS during start-up at 265VAC at full load
VDD
VOUT VDS
Ch1 Freq – 59.74kHz Ch1 Max –594V
Ch1 Max – 524V Ch2 Max – 17.1V Ch3 Max – 9.2V
(a)
(b)
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AN1715 - APPLICATION NOTE Figure 12: Output voltage ripple at 230VAC and full load
5.2. Non Isolated Flyback Description For non-isolated applications, the following solution can be used. The transformer specifications are the previous ones, but galvanic insulation and auxiliary winding are not required. The converter schematic is shown in figure 13. The VDD voltage is provided rectifying the transformer output voltage. This allows to have a supply voltage higher of 0.8V (forward voltage drop on D4), avoiding a VDD lower than VDDoff(MAX)=9V.
Figure 13: Non isolated Flyback converter
R2 AC IN
R1 10
1k
D1
BYW100-200 L1 470uH
1N4007
+10V
D4 C5 100pF
R2 150k
DZ1 9.1V
L2 22uH
D2 1N4148
C6 470uF
+
C7 47uF
+
D3 STTH106 +
C1 4.7uF 400V
+
+
C3 10uF
C2 4.7uF 400V
VIPer12A VDD
FB
C4 10nF AC IN
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TF1
DRAIN
CONTRO
SOURCE
GND OUT
AN1715 - APPLICATION NOTE Load and line regulation results are shown in figures 14 and 15 respectively, while the efficiency is shown in Figure 16. The converter shows good performance concerning the output voltage regulation, with efficiency higher than 75% at full load.
Figure 14: Load Regulation
10,6 185V 220V 265V
10,4
10,2 ) V (
T U O
V10,0
9,8
9,6 0
100
200
300
400
500
IOUT (mA)
Figure 15: Line Regulation
11,0 Rx Mode Full load
10,5
) V ( 10,0 T U O
V
9,5
9,0 180
200
220
240
260
280
VINAC (V)
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AN1715 - APPLICATION NOTE Figure 16: Efficiency Vs. output current
100 185V 230V
90
265V
80 ) % ( y 70 c n e i c i f f E 60
50
40 0
100
200
300
400
500
IOUT (mA)
In figures 17, 18 and 19 typical waveforms in RX mode and full load are shown: even in this case, in RX Mode, the converter works in burst mode, with a maximum switching frequency of 30kHz. The startup transient is shown in figure 20. The maximum drain voltage has been measured under worstcase operations, i.e. start-up @VIN=265VAC and full load. The maximum measured value is 612V, as shown in figure 20(b) and the output voltage ripple at full load and VIN=230VAC is shown in figure 21.
Figure 17: Typical waveforms at 185VAC: (a) RX and (b) Full load
Ch1 ∆Freq – 19.84kHz Ch1 Freq – 29.33kHz Ch1 Max – 380V Ch2 Max – 112mA
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Ch1 Freq – 58.47kHz Ch1 Max –454V Ch2 Max – 312mA
(a)
(b)
AN1715 - APPLICATION NOTE Figure 18: Typical waveforms at 230VAC: (a) RX mode and (b) Full load
Ch1 ∆Freq – 29.07kHz Ch1 Freq – 19.52kHz
Ch1 Freq – 58.23kHz Ch1 Max –518V Ch2 Max – 304mA
Ch1 Max – 456V Ch2 Max – 128mA
(b)
(a)
Figure 19: Typical waveforms at 265VAC: (a) RX mode and (b) Full load
Ch1 ∆Freq – 29.76kHz Ch1 Freq – 14.67kHz Ch1 Max – 512V
Ch1 Freq – 58.11kHz Ch1 Max –570V Ch2 Max – 304mA
Ch2 Max – 128mA
(a)
(b)
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AN1715 - APPLICATION NOTE Figure 20: (a) Start up time at 230VAC and (b) VDS during start-up at 265VAC at full load
VDD
VOUT VDS
Ch1 Max – 548V
Ch1 Freq – 60.11kHz
Ch2 Max – 15.2V Ch3 Max – 10.3V
Ch1 Max –612V
(a)
(b)
Figure 21: Output voltage ripple at 230VAC and full load
7. SPECTRUM FREQUENCY COMPARISON If lower switching frequency is required during RX mode due to interference issues, the primary inductance of the transformer has to be reduced down to 800µH, resulting in a higher drain peak current. This imposes a higher drain current capability device such as the VIPer22A, whose minimum peak drain current is of 560mA. In figure 22 the comparison between the solutions with VIPer12A and VIPer22A is shown: it is important
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AN1715 - APPLICATION NOTE to point out that, due to the lower primary inductance, the converter with VIPer22A in burst mode works with a lower average switching frequency respect to the solution using VIPer12A, but with a higher peak drain current. The harmonic current spectra for the two converters are shown in figure 23 and 24 respectively. It is possible to note that, in the frequency range up to 200kHz, the current harmonics amplitudes in the VIPer22A converter are lower than VIPer12A converter. This can give less interference issues when the PLMS works in RX mode.
Figure 22: Typical waveforms at 185VAC in RX Mode: (a) with VIPer12A and (b) with VIPer22A
Ch1 ∆Freq – 19.84kHz
Ch1 ∆Freq – 14.79kHz Ch1 Freq – 19.70kHz Ch1 Max – 334V
Ch1 Freq – 29.33kHz Ch1 Max – 380V Ch2 Max – 112mA
Ch2 Max – 269mA
(a)
(b)
Figure 23: Harmonic current spectrum with VIPer12A 25
VIPer12A 20
) A m15 ( e d u t i l p m10 A
5
0
0
100
200
300
400
500
Frequency (kHz)
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AN1715 - APPLICATION NOTE Figure 24: Harmonic current spectrum with VIPer22A 25
VIPer22A 20 ) A15 m ( e d u t i l p 10 m A
5
0 0
100
200
300
400
500
Frequency (kHz)
8. CONCLUSION Two solutions have been introduced in order to power a PLMS based on ST7538 chip specifications. The power supplies have been designed and developed using the VIPower device VIPer12A, since it represents the device of choice for the considered output power level in terms of performance and price. The main result of this investigation is that the proposed power supply performs well in terms of line and load regulation, working in burst mode when the PLMS works in RX mode, thus reducing the maximum switching frequency to 30kHz. Using VIPer22A it is possible to reduce the average burst switching frequency to 19KHz, since a transformer with a lower primary inductance than VIPer12A converter can be chosen. Even if this device is more expensive compared to VIPer12A, the performance in terms of frequency reduction will be improved and, consequently, the interference with the PLMS will be reduced.
Information furnished is believed to be accurate and reliable. However, STMicroelectronics assumes no responsibility for the consequences of use of such information nor for any infringement of patents or other rights of third parties which may results from its use. No license is granted by implication or otherwise under any patent or patent rights of STMicroelectronics. Specifications mentioned in this publication are subject to change without notice. This publication supersedes and replaces all information previously supplied. STMicroelectronics products are not authorized for use as critical components in life support devices or systems without express written approval of STMicroelectronics. The ST logo is a trademark of STMicroelectronics
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